A high efficiency slot array antenna with single layered gap waveguide feeding system for long-range wireless systems

Abstract

This paper presents a slot array antenna composed of H-plane horns as radiating slots and a pillbox feeding mechanism in gap waveguide (GW) technology. This new configuration presents a new full-metal slot array antenna with low complexity, fabrication cost, and high gain with directive radiation, which can make it a good candidate for use in long-range wireless systems. Making use of an H-plane reflector in the feeding part of the antenna has removed the need for a corporate feed network and enhanced the structure’s simplicity. Moreover, the use of semi-horn-shaped radiating slots has improved the antenna efficiency by resolving the grating lobe issue associated with transverse slots. These features make the proposed antenna easily scalable for higher gains without imposing extra complexity on the design and fabrication process. The GW technology also lets the proposed antenna be a promising solution for the next-generation millimeter-wave systems. In addition, the proposed fully metallic antenna leverages a design that offers high efficiency and a flat gain response over the desired bandwidth. A fabricated prototype exhibits 60–90% overall efficiency over an impedance bandwidth of about 8% ($\:{S}_{11}$ < -10 dB), and a 2.5 dB gain variation across the operating bandwidth.

Introduction

Slot array antennas are widely used in various applications such as radar, navigation, and 5G wireless systems due to their high gain, low profile, and directional radiation pattern^1,2,3. However, traditional slot array antennas suffer from limitations in bandwidth, efficiency, and manufacturability⁴, especially at mm-wave ranges. In recent years, gap waveguide technology has been introduced^5,6,7,8 and emerged as a promising solution to improve the characteristics of slot array antennas⁹¹⁰. at high-frequency bands.

Gap waveguide (GW) technology represents a significant advancement in the field of microwave engineering, particularly for applications operating within the millimeter-wave (mm-wave) frequency range. Fundamentally, this technology is predicated on the suppression of parallel plate modes between two perfect electric conductors (PEC) or perfect magnetic conductors (PMC) plates that are separated by a distance smaller than one-quarter wavelength (λ/4). This unique configuration facilitates efficient electromagnetic wave propagation while simultaneously mitigating undesirable modes that could degrade performance. One of the most compelling features of gap waveguides is their fabrication simplicity, as they can be realized without necessitating electrical contact between their metal blocks. This characteristic not only streamlines manufacturing processes but also enhances design flexibility, an essential factor in advancing mm-wave technologies where precision and adaptability are paramount. Within this domain, three principal types of GW have emerged: ridge GW (RGW), groove GW (GGW), and microstrip GW (MGW). Each variant possesses distinct structural benefits and operational characteristics that cater to specific application requirements. The introduction and evaluation of GW have paved the way for the implementation of various microwave components and antennas^{11,12,13,14,15,16,17,18,19,20,21,22}, specifically in the design of slot array antennas as a critical component in modern communication systems. To optimize antenna efficiency utilizing gap waveguide technology, various methodologies have been investigated. These include designing optimized feed networks to ensure uniform energy distribution across antenna elements, employing high-quality factor resonators to improve signal integrity, and incorporating metamaterials to manipulate electromagnetic properties effectively^{15,16,17,18,19,20,21,22,23,24}. These techniques have shown promising results in improving the efficiency and gain of slot array antennas, but still, there exist strict challenges in terms of design, setup, and manufacturing complexity.

In¹³ an 8$\:\times\:$8 slot array with a GW-based corporate feed network is designed and 16% bandwidth with 23.5 dBi gain and an efficiency of about 41% is achieved. In¹⁴ a multilayer 8$\:\times\:$8 cavity-backed slot array with a GW-based corporate feed network is developed and 12% bandwidth with an efficiency of about 65% and a maximum gain of 26 dBi over the operating frequency band is obtained. A double layer 8 × 8 PCB slot array is introduced in¹⁵ with 75% efficiency and 23.6% bandwidth. The applied corporate feed network includes cavities and RGW power dividers. In¹⁶ a multilayer 8$\:\times\:$8 slot array with a GW-based corporate feed network is presented and 26% bandwidth with 75% efficiency is obtained. In¹⁷ a wideband multilayer 8$\:\times\:$8 cavity-backed slot array with a GW-based corporate feed network is designed which introduces 30% bandwidth, a maximum gain of 27.5 dBi, and efficiency of about 80%. In¹⁸ a single layer 4$\:\times\:$4 slot array antenna fed by GW dual-mode resonators is introduced which presents 7.8% bandwidth, an average efficiency of 80%, and 18 dBi gain over the operating bandwidth. In¹⁹ a 16$\:\times\:$16-element corrugated slot antenna array with a GW-based corporate feed network is designed and 16% bandwidth with 70–80% efficiency and 32.5 dBi gain is achieved. In²⁰ a multilayer 4$\:\times\:$10 RGW slot array is designed with 3.5% bandwidth and a peak gain of 20 dBi.

All GW-based designs presented in^{13,14,15,16,17,18,19,20} introduce significant complexity in both the design and fabrication process due to the rather complex structure of the corporate feed network. Moreover, as the antenna aperture size increases, the insertion loss increases as well which can lead to a significant reduction in antenna efficiency. To resolve this problem, series-fed long transverse slot arrays have been applied^{23,24,25,26,27}. However, narrow band performance and low efficiency due to the grating lobe problem are the rising challenges to deal with. In^{23,24,25,26,27} substrate-based topologies have been introduced to shrink the guided wavelength inside the feeding waveguide, resulting in grating lobe suppression. However, making use of substrate as the wave path leads to a reduction of antenna efficiency, especially at mm-wave ranges, along with lower power handling capability at lower frequencies.

In this paper, a proof-of-concept is presented for a new full metal slot array antenna which offers a very simple structure with low design and fabrication complexity even for large aperture sizes. The proposed structure employs GW technology along with the H-plane horn-like long slots and a symmetric parabolic H-plane reflector as the feeding system. Making use of an H-plane reflector in the feeding part of the antenna has removed the need for a corporate feed network and enhanced the structure’s simplicity. Moreover, the use of semi-horn-shaped radiating slots has improved the antenna efficiency by resolving the grating lobe issue associated with transverse slots. These features make the proposed antenna easily scalable for higher gains without imposing extra complexity on the design and fabrication process. The designed antenna prototype exhibits 8% impedance bandwidth (15.7–16.9 GHz), a peak gain of 24.5 dBi, and 70–90% efficiency over most of the operating bandwidth. The presented structure, due to its high gain and directional radiation pattern, can be applied in long-range wireless systems such as radar, navigation, and point-to-point communication systems. Moreover, this topology can be a good candidate for use in 5G and mm-wave wireless systems as it provides the advantages of GW technology along with high gain and efficiency.

The following sections present the topology and the design rules of the proposed structure, including the feeding system and the semi-horn radiating slots. Subsequently, the simulation and measurement results of the designed prototype are presented and discussed. Finally, concluding remarks are provided.

Antenna structure and design

The proposed antenna topology, as illustrated in Fig. 1, represents a significant advancement in the design of high-frequency antennas. This innovative structure is composed of a GW-based pillbox that serves as the feeding section and is intricately connected to an array of series-fed long slots functioning as the radiating element. The configuration further incorporates a two-dimensional parabolic reflector and a parallel plate waveguide, which are strategically enclosed by a bed of nails and topped with a metal lid separated by an air gap denoted as “g”. Within this sophisticated design, each pin exhibits a square cross-section characterized by specific dimensions: “b” signifies the width of each pin, while “d” denotes its height. The spacing between individual pins is referred to as “a”, which plays a crucial role in optimizing performance. The careful selection of these parameters, specifically “g”, “d”, “a”, and “b” (as shown in Fig. 2) according to the “Fakir bed of nails” design principles facilitates the achievement of an effective stopband for parallel plate modes within the targeted frequency range⁷. Figure 2 complements this discussion by depicting the dispersion diagram corresponding to the designed pin surface; it clearly illustrates a stopband extending from 10.5 GHz to 20.6 GHz, encompassing the entire Ku-band (12–18 GHz). The dispersion diagram shown in Fig. 2 illustrates the normalized propagation constant (β/k₀) versus ‎frequency for the first four parallel plate modes between the periodic pin surface and the ‎upper plate. The fundamental mode (Mode 1) is the quasi-TEM mode of the parallel plate ‎waveguide. The stopband, extending from 10.5 GHz to 20.6 GHz, is characterized by the ‎condition β/k₀ ≈ 0 for Mode 1. Within this bandgap, the pin surface exhibits a high impedance surface ‎ ‎(PMC-like behavior), suppressing the propagation of all parallel-plate modes inside the gap. ‎Modes 2–4 represent higher-order Bloch modes of the parallel plate waveguide between the perforated surface and the upper plate. The existence of the stopband is the fundamental principle ‎that enables the low-loss operation of the GW technology.

Feeding part

As mentioned above, the feeding section of the proposed antenna is composed of a pillbox and a parallel plate waveguide connected. In the pillbox section, a probe of height “d₁” is positioned at the parabolic reflector’s focal point (see Fig. 3). As shown in Fig. 3, the focal length of the parabolic reflector is denoted by “f”. Due to the symmetric configuration of the reflector, the probe is positioned along the parabola’s axis. The probe generates a cylindrical wave toward the reflector wall, which subsequently converts it into a plane wave after reflection. Similar to the design presented in²⁸, a pin, we call it a corrective pin (see Figs. 1 and 3), with the diameter “W” and the height of “d2” positioned by the feeding probe. This pin acts as a very simple sub-reflector that suppresses the intersection between the cylindrical and plane waves in the parallel plate waveguide area which leads to the generation of a high-quality and pure plane wave propagating inside the parallel plate waveguide section. At the same time, this pin introduces a blockage at the center of the reflector aperture. So, its diameter should be carefully optimized to minimize its blockage effect and maximize its corrective impact on the aperture illumination. The influence of the corrective pin on the field distribution within the feeding section is investigated, and the results are depicted in Figs. 3(b, c). In Fig. 3(b), we see that for the case without the corrective pin, the propagating plane wave inside the parallel plate waveguide is not of good quality, as it is a mixture of the reflected plane wave from the reflector and the incident cylindrical wave originating from the probe and propagating toward the parallel plate waveguide. As a result, as shown in Fig. 3(b), the amplitude uniformity of the outgoing plane wave is deteriorated. However, in the case of the corrective pin (Fig. 3(c)), the outgoing plane wave amplitude uniformity is enhanced as the corrective pin effectively suppresses the probe from directly radiating toward the parallel plate waveguide. We should note that in this simulation, to see only the feeding section performance, the radiating slots do not exist, and the parallel plate waveguide is an open circuit, so that the propagating wave inside it can radiate through its open end.

Fig. 1

The 3D expanded view of the proposed antenna geometry. (Generated by ANSYS HFSS software, version 15.0.3, http://www.ansys.com/products/electronics/ansys-hfss).

Fig. 2

Dispersion diagram of the designed unit cell of the pin surface in an infinite environment.

Fig. 3

(a) The geometrical parameters of the corrective pin in the GW-based pillbox structure (Generated by ANSYS HFSS software, version 15.0.3, http://www.ansys.com/products/electronics/ansys-hfss), (b) the simulated field distribution inside the feeding part without a corrective pin, and (c) with a corrective pin.

Radiating layer

As shown in Fig. 1, the radiating aperture of the antenna comprises a 1$\:\times\:$4 array of H-plane horn-like slots positioned transversely on the parallel plate waveguide’s top plate. The radiating elements are fed in series by a plane wave propagating within the parallel plate waveguide. The subsequent sections delve into the detailed design of each radiating slot, representing the array unit cell, and the overall array configuration.

Unit cell

Figure 4 illustrates the simulation model for an individual radiating slot/array unit cell. The figure also details each slot’s geometric parameters. To model an infinite array environment, periodic boundary conditions (PBC) are applied as shown in Fig. 4. The PBCs in the x-direction model the long radiating elements while those in the y-direction model an array of radiating elements spaced by “Ps” in the y-direction. This approach accounts for mutual coupling between array elements in the E-plane (z-y plane). Figure 5(a) shows the simulated E-plane radiation patterns of both the unit cell and the 1 × 4 array using the pre-mentioned simulation model. We see that by making use of the proposed unit cell the grating lobe problem is not raised. In Fig. 5(b), the S-parameters, including S11 and S12, of the array unit cell are also depicted. The dimensions of the unit cell used in this simulation can be found in Table 2, where Ws = 8 mm. The radiation efficiency of the radiating slot can be extracted by making use of the achieved S-parameters, which are discussed in the following section.

The dimensions of the stairs used in the geometry of the slots, as shown in Fig. 4, significantly influence the reflection coefficient introduced by each radiating element. The width of the transverse slot “Ws” controls the radiation power of each element. The radiation efficiency of each element can be calculated using the unit cell simulation model shown in Fig. 4 as done in the following section.

By carefully selecting “Ws” for each radiating element, we can achieve the desired E-plane aperture illumination, whether tapered or uniform. This choice directly impacts the side lobe level (SLL) and directivity. Here, we aim for uniform aperture illumination to maximize aperture efficiency and directivity.

The length of the radiating H-plane horns, which controls the H-plane beamwidth, is effectively determined by the feeding GW-based pillbox aperture width (“D” in Fig. 3). However, the width of the radiating horns (“Ws₂” in Fig. 4) is constrained by the element spacing (“Ps” in Fig. 4). The spacing between adjacent transverse slots should be equal to the free space wavelength (λ₀) to provide in-phase excitation for the whole array and consequently a broadside radiation pattern.

Fig. 4

The simulation model for the array unit cell, including its geometrical parameters (a) 3D view (Generated by ANSYS HFSS software, version 15.0.3, http://www.ansys.com/products/electronics/ansys-hfss and captioned by Microsoft VISIO) (b) top view (c) side view.

Fig. 5

Simulation results obtained by the model illustrated in Fig. 4, (a) The E-plane radiation patterns of both the unit cell and the 1$\:\times\:$4 array (b) S-parameters of the unit cell.

Array

As shown in Fig. 1, a 1$\:\times\:$4 array of the long slots shown in Fig. 4 is used as the antenna aperture. The radiating slots are transversely fed in series by a plane wave generated in the feeding section. To excite all the slots in-phase, the spacing between adjacent slots has to be equal to λ₀. However, this fact raises the grating lobe issue caused by the array factor. So, to deal with the grating lobe problem, as shown in Fig. 4, the slot width “Ws” is enhanced to “Ws2” using a horn-like taper to correct the unit cell radiation pattern. The number of radiating slots, which influences the E-plane beamwidth, is restricted by the desired 3 dB gain bandwidth. Increasing the number of elements can lead to reduced 3 dB gain bandwidth due to the long-line effect and beam squint.

The model illustrated in Fig. 6 presents a comprehensive analysis of the forward wave as it traverses through the feeding waveguide. However, this analysis must also account for the presence of backward waves that arise due to reflections from the short end of the waveguide. These reflections not only contribute to overall system performance but also significantly influence radiative properties. As depicted, each radiating element within the configuration operates under dual conditions, while one serves as a terminal for forward wave propagation, it concurrently functions as an initiator for backward wave radiation. This interdependence suggests that there exists a reciprocal relationship between these two phenomena, specifically, the residual power emanating from the forward wave denoted as Pout(n) in Fig. 6, effectively acts as input power for its corresponding backward counterpart. To thoroughly evaluate this system’s efficiency and output characteristics, one can derive an expression for total radiated power from each radiating slot by integrating contributions from both forward and backward waves into consideration. According to Fig. 6, the radiation efficiency of each radiating slot “η” can be computed using (1) in which the S-parameters are obtained by making use of the simulation model presented in Fig. 4. In this way, for the ith horn slot with specified Ws, one can compute η_i. Then the radiated power through each slot “P_rad(i)” can be computed making use of (2) and (3).

$$\eta=1-\:{\left|{S}_{11}\right|}^{2}-{\left|{S}_{21}\right|}^{2}$$

(1)

 

$$\:{P}_{rad\left(i\right)}={\eta\:}_{i}{P}_{out(i-1)}$$

(2)

$$\:{P}_{out\left(i\right)}={(1-\eta\:}_{i}){P}_{out(i-1)}$$

(3)

Fig. 6

The radiation model for the forward wave propagating through the feeding waveguide.

Figure 7 illustrates the “η” values associated with a radiating slot using the dimensions listed in Tables 1 and 2. It should be noted that all the slots have the same dimensions except for “Ws”. In Table 1, the “Ws” value for each radiating slot is shown. In Fig. 8, the relative radiated power through the slots is depicted and we see quite a uniform amplitude distribution across the aperture in the E-plane.

Fig. 7

The simulated “η” of a single radiating slot for different values of “Ws”.

Table 1 The width “Ws” for each radiating slot in the designed antenna.

Fig. 8

The radiated power through the slots at 16.4 GHz. The term “Prad_forward” refers to the power radiated by the forward wave and “Prad_Total” indicates the total radiated power associated with both forward and backward waves.

To enhance the efficiency of the proposed structure, we made a semi-circular taper between the edges of adjacent slots as shown in Fig. 9. In this way, we can slightly increase between the slot edges which can lead to lower mutual coupling and consequently higher efficiency. In Fig. 10, the effect of the introduced taper between slots on the antenna gain is shown. We see that this taper has provided about 0.5dBi gain enhancement.

Fig. 9

Proposed slot array antenna (a) without and (b) with semi-circular taper between slot edges.

Fig. 10

A comparison of the antenna gain for the two cases is shown in Fig. 9.

Table 2 Geometrical parameters of the designed prototype.

Results and discussion

According to the descriptions presented in the previous sections, a sample of the proposed antenna has been designed to operate in the Ku band. The dimensions of the designed antenna are listed in Tables 1 and 2. The designed prototype is fabricated (see Fig. 11) and measured in an anechoic chamber. It is also simulated using HFSS. The simulation and measurement results are illustrated in Figs. 12, 13, 14, 15 and 16.

Figure 12 shows the simulated and measured reflection coefficients, and both of them indicate proper impedance matching ($\:{S}_{11}$ < −10 dB) over the 15.7–17 GHz frequency range which shows the impedance bandwidth of 8%. In Fig. 13, the simulated E-field distribution inside the designed antenna in the case where the radiating slots exist, is depicted and a quite uniform amplitude distribution can be observed in the E-plane. Figures 14 and 15 show the E- and H-plane radiation patterns, respectively, at three frequencies over the antenna operating bandwidth. A good agreement between the simulated and measured results can be observed in a 25 dB range. We have to note that as the dynamic range of our available anechoic chamber is limited to 25 dB, there is no measured data for radiation patterns beyond the 25 dB range to show here. In Figs. 14 and 15, we see a broadside fixed beam with proper radiation performance over the operating bandwidth which reveals the uniform phase excitation of the radiating slots. Furthermore, the side lobe level of about − 13 dB in E-plane patterns confirms the quite uniform amplitude excitation of the radiating slots. The antenna gain along the boresight direction is also depicted in Fig. 16, and we see that both the measured and simulated results exhibit similar trends. The measured gain ranges from 22 to 24.5 dBi over the 15.7–17 GHz which shows a quite flat gain response over the operating frequency range. In Fig. 17, the efficiency characteristics of the presented antenna are also illustrated. The measured and simulated overall efficiencies are calculated by subtracting the measured and simulated realized gains from the maximum available directivity, respectively. In this way, we computed the antenna’s overall efficiency, which better describes the antenna radiation performance in real applications. In this way, all losses due to the ohmic losses, mismatch factor, and antenna aperture are taken into consideration simultaneously. In the measured result, even the loss due to the polarization mismatch is also taken into consideration²⁹. We see that both measurement and simulation results show the overall efficiency of more than 60% over most of the operating bandwidth, with the maximum value of about 90% which shows quite high efficiency performance for the proposed configuration.

Fig. 11

Fabricated prototype (a) front & side view (b) bottom and top layers (c) bottom view showing the SMA connector for RF feed (d) 3-D view with labeled overall dimensions (“Lt” =242 mm, “Wt” =200 mm, and “ht” =24 mm).

Fig. 12

Simulated and measured reflection coefficients of the designed antenna.

Fig. 13

The simulated E-field amplitude distribution inside the designed antenna.

Fig. 14

E-plane radiation patterns of the designed antenna at (a) 15.9 GHz (b) 16.3 GHz (c) 16.7 GHz.

Fig. 15

H-plane radiation patterns of the designed antenna at (a) 15.9 GHz (b) 16.3 GHz (c) 16.7 GHz.

Fig. 16

Simulated and measured gains of the designed antenna.

In Table 3, a comparison is provided between the proposed slot array antenna and several works on slot array antennas with broadside radiation reported in the literature. We see that the proposed antenna presented in this paper demonstrates higher efficiency and wider operating bandwidth compared to²³ and²⁷, the works in which a plane wave is used to excite long transverse slots. The antennas presented in²³ and²⁷ are substrate-based and suffer from dielectric loss, so they provide more losses and lower power handling capability compared to the proposed full-metal antenna in this paper. Furthermore, the presented antenna here introduces lower fabrication complexity especially for large apertures compared to those in which a corporate feed network (CFN) is applied. The proposed topology, compared to the presented CFN-based topologies, i.e. the ones introduced in^{13,16,17,19,21,22}, may also introduce lower loss when higher gain values and larger apertures are required. The reason is that in the proposed antenna the redesign procedure to obtain higher gain includes the variation in the reflector aperture size and the number of slots and does not significantly affect the wave path geometry in the feeding part, while in CFN-based topologies the overall CFN needs to be redesigned and enlarged which significantly affects the wave path geometry and length inside the feed network leading to considerable enhancement of losses introduced by the feed network.

Fig. 17

Simulated and measured overall efficiencies of the designed antenna.

Table 3 Comparison between the proposed antenna‎ and some reported GW-based slot array antennas.

Conclusion

This paper introduces a novel slot array antenna characterized by H-plane horn-shaped slots, which are fed in series by a plane wave generated via a GW pillbox. The innovative design not only simplifies the configuration but also enhances the overall performance of the antenna, achieving remarkable efficiency levels. The proposed structure demonstrates an impressive maximum efficiency of approximately 90% at a frequency of 16.4 GHz. This high level of efficiency is attributed to the synergistic interaction between the pillbox feeding system and the horn-shaped radiating elements. The use of this feeding mechanism minimizes losses typically associated with conventional antennas, thereby maximizing power transfer and improving radiation characteristics. A noteworthy aspect of this design is its ability to deliver consistent radiation performance across its operational bandwidth. Specifically, it exhibits a flat gain response while maintaining an efficiency range between 70% and 90% over an 8% bandwidth. Such stability in performance is crucial for various applications that demand reliable signal transmission and reception. In summary, this study presents significant advancements in slot array antenna technology through the introduction of H-plane horn-shaped slots fed by a GW-based pillbox. With its combination of high gain, simplicity in configuration, and robust radiation performance across bandwidths, this new antenna structure stands poised to contribute meaningfully to both theoretical research and practical applications within telecommunications and related fields.