Introduction

The upcoming wireless communications system is the mm-wave fifth-generation (5G) technology. Because of the exponentially growing demands of high data rates with low latency and inadequate BW in the microwave frequency (sub-6 GHz), the upcoming advanced communication system will be implemented in mm-wave frequency region mainly from 24 to 100 GHz1. The 5G mm-wave communications are expected to revolutionize daily life through its excellent connectivity with modern technology like smart factories, automotive cars, smart homes, telemedicine, virtual reality, etc.2,3,4,5. Standardization organizations, telecom companies, and government authorities are working on deploying and standardizing mm-wave 5G communications systems. The 26 GHz, 28 GHz, 37 GHz, and 38 GHz frequencies are allocated by different countries for 5G new radio (NR)6,7. As a result, antenna development in mm-wave frequency has gained significant importance in recent years8. Patch antennas are attractive for their low profile, low cost, and planar structure, but they have disadvantages such as low gain and narrow bandwidth9,10,11. The planar antenna with high gain in communication systems is attractive for satellite or base station applications. The upcoming 5G wireless communication systems will be implemented in the mm-wave frequency, but these frequencies go through high free space path loss. Hence, the communication range degrades than the 4G (fourth generation) frequency band. High-gain antennas are required on both ends of the communication device to overcome atmospheric and high propagation loss in mm-wave frequency12,13. From this perspective, the metamaterial is a strong candidate for designing an antenna with this required feature14,15,16,17,18. Various works were done to increase the antenna performance, such as gain and bandwidth. Usually, the metamaterial is positioned on the antenna radiating element or behind the defected ground antenna19, where the gap between the antenna and meta-surface layer increases the bulkiness and complexity of the antenna. An antenna coupled with a metamaterial superstrate, where a lens is positioned on the radiating patch, can enhance the gain by focusing the radiation beam. This type of antenna can provide wideband operational frequency but suffers from low radiation efficiency because of its bulky size and losses in the thick dielectric material. In research20,21,22, the antenna gain significantly increases by arranging the metamaterial superstrate on the antenna patch, but this technique has mechanical complexity and high-profile properties because of the air cavity among the superstrate and antenna. Similarly, a metamaterial structure is vertically placed to increase the gain of the bow-tie antenna23. In24,25, metamaterial was vertically positioned in the direction of the EM wave propagation to enhance the gain. The meta-surface without any air gap can improve the performance of the planar antenna by utilizing the multiple layers technique, but this can increase the design complexity of the antenna. Also, this arrangement has some major disadvantages, such as bulkiness, narrowband, and increased fabrication cost. Alternately, the antenna array configurations can achieve a high gain, but they provide a similar channel capacity to a single antenna because the array antenna is fed by a single port like a single antenna and also suffers from power loss in complex power divider networks26. On the other hand, meta-surface was generally used with patch and slot antennas because of their controlling property of EM waves6. Single-layer antennas with metamaterial were presented in20,21,22,23, where most are in the microwave range and have low gain or bandwidth. However, the MIMO performance of these antennas was not investigated. The MIMO performance analysis is crucial for this mm-wave frequency range because the MIMO antenna will provide multiple channels for increasing the data traffic, spectral efficiency, link reliability and capacity27,28,29. Hence, the antenna with a MIMO configuration is the key requirement for mm-wave 5G applications30. After the above discussions, a high-gain antenna with a larger operational bandwidth is highly recommended for mm-wave 5G applications; moreover, MIMO performance analysis is also essential.

This paper presents a metamaterial incorporated planar dipole antenna for mm-wave 5G applications. The designed HRIMDA gain is increased by 3.5 dBi, using five series of horizontal H-shape metamaterial structures and operational bandwidth from 23 to 38 GHz. The investigation of the MIMO configurations of HRIMDA shows the high isolation among two conjugated antennas, which is < − 30 dB. The MIMO parameters like ECC and DG values are < 0.0001 and > 9.99, respectively, which are the near-ideal values. Finally, the proposed HRIMDA has a high gain and a more comprehensive operational bandwidth at mm-wave frequency. Moreover, the MIMO configuration shows excellent parametric behaviour, which makes the planned antenna attractive for mm-wave 5G applications.

Metamaterial unit cell design

Metamaterials (MTMs) are commonly referred to as artificial materials with unique, effective medium characteristics that are not generally found in nature. These unique properties, like negative permittivity and permeability, gained significant attention in various applications like antenna design, clocking, and sensing31,32,33. The electric and magnetic quantities, also known as essential parameters, explain a medium’s EM wave behaviour. This article presents a horizontal H-shaped metamaterial resonator for gain enhancement over the more extensive mm-wave frequency range. The proposed horizontal H shape metamaterial is designed on the Rogers RT5880 substrate material with a thin 0.254 mm thickness. The parametric sketch is presented in Fig. 1a, where a = 1.6 nm, r = 1.10 nm, w = 0.10 nm, and g = 0.50 nm, respectively. The simulation setup of the designed MTM is presented in Fig. 1b, where a perfect electric conductor (PEC) and perfect magnetic conductor (PMC) are applied along the y and z axes, respectively. The EM wave is applied along the x-axis in a positive direction. The simulation of the proposed MTM is done by the CST microwave studio34.

Fig. 1
figure 1

Horizontal H shape metamaterial unit cell (MUC) (a) front view and (b) simulation setup.

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The MTM electromagnetic property can be characterized by the Lorentz-Drude model. The Eqs. (1–3) are used to calculate the relative permittivity (ɛr), relative permeability (µr) and effective refractive index (ηr),35,36. Where, S11 and S21 are the reflection and transmission coefficients, respectively. d is the propagation distance of the medium, and Ko is the wave number.

$$\varepsilon_{r} = {\raise0.7ex\hbox{$2$} \!\mathord{\left/ {\vphantom {2 {jk_{0} d}}}\right.\kern-0pt} \!\lower0.7ex\hbox{${jk_{0} d}$}} \times {\raise0.7ex\hbox{${\left( {1 - S_{11} - S_{21} } \right)}$} \!\mathord{\left/ {\vphantom {{\left( {1 - S_{11} - S_{21} } \right)} {\left( {1 + S_{11} + S_{21} } \right)}}}\right.\kern-0pt} \!\lower0.7ex\hbox{${\left( {1 + S_{11} + S_{21} } \right)}$}}$$
(1)
$$\mu_{r} = {\raise0.7ex\hbox{$2$} \!\mathord{\left/ {\vphantom {2 {jk_{0} d}}}\right.\kern-0pt} \!\lower0.7ex\hbox{${jk_{0} d}$}} \times {\raise0.7ex\hbox{${\left( {1 - S_{21} + S_{11} } \right)}$} \!\mathord{\left/ {\vphantom {{\left( {1 - S_{21} + S_{11} } \right)} {\left( {1 + S_{21} - S_{11} } \right)}}}\right.\kern-0pt} \!\lower0.7ex\hbox{${\left( {1 + S_{21} - S_{11} } \right)}$}}$$
(2)
$$\eta = \sqrt {\varepsilon_{r} \mu_{r} }$$
(3)

Figure 2 demonstrates the EM parameters property of the intended horizontal-H shape MUC, which shows permittivity near zero (ENZ) value at the upper-frequency region. The permeability near zero (MNZ) has appeared from the lower operating frequencies to 35 GHz. This permittivity and permeability value leads to a higher index metamaterial. The real value of the refractive index shows a higher refractive index metamaterial (HRIM) property than the air medium. This metamaterial property is used to enhance the antenna’s performance in terms of gain and directivity37 and provides a unique way to control the direction of the EM radiation based on Snell’s law. Figure 3 presents the EM wave propagation from air to HRIM and HRIM to air. According to Snell’s law, when an incident angle \({\theta }_{i}\) fixed the refracted angle \({\theta }_{r}\) shows a smaller value for EM wave propagation through a low-index to high-index medium. On the other hand, for the same conditions, the EM wave shows a larger refracted angle when propagating through the high index to the low-index medium. Based on this, a metamaterial lens can be formed and can be focused on the EM wave in the direction of propagation and aggregate the beam towards the end-fire direction38. The energy transmitted through metamaterial congregates and is intense in the direction of propagation. This type of MTM can enhance the gain and directivity of the antenna. The EM wave aggregation property can be understood from the metamaterial’s electric (E) field39, where a spherical wave converts to a planar wave and decreases the half-power beamwidth by metamaterial. This process was implemented by various metamaterial structures on planar antenna39,40.

Fig. 2
figure 2

Metamaterial property of the designed horizontal H shape MUC.

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Fig. 3
figure 3

EM wave propagation through designed HRIM.

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The characteristics of the metamaterial are analyzed by surface current allocation, E-field, and H-field distribution, shown in Fig. 4. The relationship between these three field distributions can be understood by analyzing Maxwell’s equations41,42, where the electric (\(\vec{D}\)) and magnetic field induction (\(\vec{B}\)) can be described by Eqs. 4 and 5. The ɛₒ and µₒ represent the permittivity and permeability of free space ɛr, and µr represents relative permittivity and permeability.

$$\vec{D} = \varepsilon \vec{E} = \varepsilon_{o} \varepsilon_{r} \vec{E}$$
(4)
$$\vec{B} = \mu \vec{H} = \mu_{o} \mu_{r} \vec{H}$$
(5)
Fig. 4
figure 4

(a) E-field. (b) H-field and (c) surface current distribution of the designed MUC.

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Figure 4 shows the E-field allocations of the intended MUC for 24 GHz, 28 GHz, 30 GHz, and 37 GHz frequencies, which shows the high E-field density at the left and right edges of the MUC. Also, the field intensity proportionally increases with frequency. This high E-field medium aggregates the incident EM wave to the propagation direction. The low magnetic field distribution of the designed MUC is presented in Fig. 4. The magnetic field intensity of the upper side of the metamaterial increases with the frequency, and the lower side of the MUC proportionally decreases. The surface current allocation of the MUC is presented in Fig. 4, where anti-parallel current flow appeared at the splits of the MUC. The surface current intensity gradually decreases with the increment of the frequency. This property makes EM wave aggregation capability higher at the upper-frequency region than at the lower frequency.

Broadband dipole antenna design and metamaterial incorporation

Single antenna design

The design configuration of the proposed broadband dipole antenna (BDA) is depicted in Fig. 5. The antennas were designed on Rogers RT5880 substrate material with a dielectric constant of 2.2, a loss tangent of 0.0009, and a substrate thickness of 0.254 mm. The low dielectric constant is chosen for its ability to reduce electrical power loss at the higher frequency application43. On the other hand, low substrate thickness would decrease the end-fire cross-polarisation44. The substrate feature improves the antenna results parameters at the higher frequency regions. The designed antenna is fed by a W4 = 0.71 mm feed line, which is a series with W3 = 0.5 mm and a 0.3 mm width inverted U-shaped structure. All gematrical sketches of the intended antenna are presented in Fig. 5, and the design’s parameter values are listed in Table 1.

Fig. 5
figure 5

Front view of a designed BDA with design parametric sketch.

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Table 1 Design parameters.
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The return loss (S11), efficiency, and gain plot of the designed antenna are presented in Fig. 6. This demonstrates the realized gain of the designed antenna between the range of 4.7 to 6 dBi at the resonant bandwidth (24.35–38.7 GHz). The maximum 6 dBi gain appeared at the 34 GHz frequency. Conversely, the efficiency shows around 95% at the resonant bandwidth. The intended antenna radiation pattern at 26 GHz and 28 GHz frequencies in the E-plane and H-plane is presented in Fig. 7. The pattern shows good directivity of the designed BDA. The cross-polarisation is lower than 8 dB, and front to the back ratio of co-polarisation is more than 15 dB for the entire operating frequency. Figure 8 presents the designed antenna’s electric (E) and magnetic (H) field distribution. The E-field intensity of the dipole arm is higher at 28 GHz frequency than at 26 GHz, which indicates the higher directivity and gain at 28 GHz frequency.

Fig. 6
figure 6

Efficiency and gain plot of the designed BDA.

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Fig. 7
figure 7

simulated radiation pattern of the designed BDA.

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Fig. 8
figure 8

Magnetic (H)-field at (a) 26 GHz, (b) 28 GHz and electric (E) field (c) 26 GHz, (d) 28 GHz of the designed BDA.

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Figure 9 shows the bandwidth (S11) plot of the different designs. In design one, the BDA does not show any resonance because of the incompetence of the feeding. The feeding on the dipole is completed in design two by making a slot line from the middle of the dipole to the inverted U shape. This design shows 5.54 GHz (− 10 dB) bandwidth from 30 to 35.54 GHz. The antenna’s bandwidth is further enhanced in design 3 by making an elliptical slot at the center of the inverted U-shaped line, which increases the − 10 dB resonance bandwidth (10.8 GHz) from 28 to 38.8 GHz. Finally, a broadband 14.35 GHz (− 10 dB) bandwidth is achieved from 24.35 to 38.7 GHz frequency by declining the dipole arm by 40-degree.

Fig. 9
figure 9

(a) design evaluation of the BDA. (b) reflection coefficient (S11) of the different steps of antenna designs.

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Metamaterial incorporated antenna

The BDA’s gain is increased significantly by incorporating the metamaterial structure into the propagation direction of the EM wave. Figure 10 shows the gradual incorporation of HRIM. The S11 of the different HIRM stages is outlined in Fig. 11a, which shows a stable − 10 dB impedance bandwidth. The operational bandwidth of the final 5 × 6 HRIM appeared from 23 to 38 GHz frequency. The HRIM’s effect on the BDA’s gain is illustrated in Fig. 11b. The BDA’s gain increases gradually with the increase of the HRIM unit cell series, and for the final 5 × 6 HRIM base antenna, the gain is achieved from 6 to 9.5 dBi. The highest gain of the antenna reached 9.5 dBi at 35 GHz resonant frequency. The equivalent circuit (EC) of the proposed HRIMDA is presented in Fig. 11c based on45,46,47, where LC represent the feed line, R1L1C1 and R2L2C2 represent dipole antenna and R3L3C3 represent HIRM. The EC is simulated by Advanced Design System (ADS) software, which agrees well with CST simulation shown in Fig. 11d. The EM wave aggregation property of the proposed HRIMDA can be understood from the antenna’s electric, magnetic, and power flow. Figure 12 shows the proposed electric and magnetic fields at 26 and 28 GHz frequencies. The HRIM unit cell aggregates the EM wave in the direction of propagation. This EM wave aggregation engineering can be more clarified by the power flow of the HRIMDA, which is depicted in Fig. 13. The normal and HRIMDA power flow is presented to understand this behaviour. The circled region in Fig. 13 shows the power flow at the side edge of the antenna significantly reduced by the incorporation of HRIM on the antenna. The power flow significantly boosts the direction of the EM wave propagation.

Fig. 10
figure 10

Incorporation of the HRIM series on the BDA.

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Fig. 11
figure 11

(a) S11 and (b) gain plot of the intended HRIMDA at a different stage, (c) EC model (L = 1.029 nH C = 0.0259 pF, R1 = 3.493 Ω, L1 = 1.814 nH, C1 = 0.00872 pF, R2 = 95.011 Ω, L2 = 0.660 nH, C2 = 0.032 pF, R3 = 3.27 Ω, L3 = 4.550 nH, C3 = 0.010 pF and (d) S11 of ADS and CST simulation.

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Fig. 12
figure 12

(a) H-field distribution and (b) E-field distribution of the proposed HRIMDA.

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Fig. 13
figure 13

Power flow at 28 GHz frequency (a) normal BDA and (b) proposed HRIMDA.

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Fabrications and experimental results

To verify the intended HRIMDA’s gain enhancement, two antennas (typical BDA and HRIMDA) were fabricated using 2.54 mm thin Rogers RT5880 (ɛr = 2.2) substrate material. The size of the antenna is 10 × 18 mm2. Figure 14 shows the front and rear sides of the fabricated typical and HRIMDA. The vector network analyzer and far-field anechoic chamber measurement systems measured the fabricated antennas, as shown in Fig. 15. The simulated and measured S11 of the typical and HRIMDA is plotted in Fig. 16a, and the S11 of the fabricated antenna agrees well with the simulated one. The measured bandwidth of the HRIMDA is 23–38 GHz frequency. The measured gin of the normal and HRIMD is presented in Fig. 16b, which shows good agreement with the simulation results.

Fig. 14
figure 14

Fabricated typical BDA (a) front view (b) rearview; and Fabricated HRIMDA (c) front view (b) rearview.

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Fig. 15
figure 15

Anechoic chamber measurement system.

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Fig. 16
figure 16

(a) Simulated and measured S11, and (b) Simulated and measured gain of the typical dipole and HRIMDA.

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The simulated and measured E-plane and H-plane radiation patterns at 26 GHz and 28 GHz of the proposed HRIMDA are presented in Fig. 17. The measured radiation pattern agrees well with the simulated one. The cross-polarisation is less than 10 dB, and front to back ratio of the co-polarisation is less than 18 dB at the entire operating frequency, indicating the radiation beam’s directivity. A detailed comparison of the proposed antenna with the existing broadband mm-wave antennas is presented in Table 2. The defective ground antenna is presented in48,49, where wideband is achieved by defective ground technique but shows low antenna gain. Different planar-printed log-periodic dipole array (PLPDA) antenna types were presented in38,39,50. For broadband 5G mm-wave applications, the different types of MTM structures showed significant gains. A planar patch antenna6, Shared aperture antenna44, Bow-tie antenna12 and Antipodal Vivaldi antenna51 with meta-surface were presented for 5G applications, where the gain was improved significantly by utilizing different MTM structures. Multiple-layer microstrip antenna22 and dual-layer two-element slotted patch antenna52 are also presented, where MTM was placed on top of the antenna with an air gap. The MIMO parameters analyses were performed only in6,22,52. Finally, the proposed planar HRIMDA achieved a significant percentage bandwidth enhancement (BW %) compared to the other listed antenna. Also, HRIM enhances gain by 3.5 dBi, which is higher than the existing antennas. The investigation of the MIMO parameters analysis, compact antenna size, larger BW %, and higher gain prioritize the proposed antenna over the present antennas.

Fig. 17
figure 17

Simulated and measured normalized radiation pattern of the intended HRIMDA.

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Table 2 Comparison with the existing broadband mm-wave antenna.
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MMO configurations

Figure 18 shows a four-element MIMO antenna arrangement of the intended HRIMDA. The designed MIMO configuration is fabricated, and isolation of the two adjacent MIMO elements (antenna 2 and 3) is measured. The front and back views of the fabricated antenna are depicted in Fig. 19. Figure 20a shows the S11 of the simulated and measured MIMO elements. The simulated and measured isolation of the MIMO elements is depicted in Fig. 20b and c, where the measurement results agree well with the simulated results.

Fig. 18
figure 18

Four-element MIMO antenna system.

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Fig. 19
figure 19

Fabricated MIMO antenna (a) front view and (b) back view.

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Fig. 20
figure 20

Simulated and measured (a) S11 of the MIMO element, (b) Simulated isolation between the MIMO element, (c) Measured isolation between the MIMO element, and (d) MIMO parameters of two conjugate antennas (antenna 2 and 3).

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The performance analysis of the designed MIMO antenna is done by investigating the envelop correlation coefficient (ECC) and diversity gain (DG). Equation 6 calculates the ECC value, which indicates the correlation of the MIMO antenna element. Equation 7 calculates the DG value, which indicates the impact of the antenna’s radiated power diversity scheme.

$$\rho_{ec} (i,j) = \frac{{\left| {S_{ii} \times S_{ij} + S_{ji} \times S_{jj} } \right|^{2} }}{{\left( {1 - \left| {S_{ii} } \right|^{2} - S_{ij}^{2} } \right)\left( {1 - \left| {S_{ji} } \right|^{2} - S_{jj}^{2} } \right)}}$$
(6)
$$DG = 10 \times \sqrt {1 - \rho_{ec} }$$
(7)

In Eq. 6, Sii and Sjj are the reflection coefficients; Sij and Sji (i ≠ j) are the transmission coefficients of MIMO elements. The ECC value of the MIMO antenna should be less than 0.1, and a lower ECC value improves the MIMO performance. The simulated and measured ECC value of two conjugate antenna elements presented in Fig. 20d shows an ECC of less than < 1 × 10–3, which is the nearly ideal value. The simulated and measured DG value is achieved above 9.9, where the ideal value is 10. A comparison of the proposed four-element MIMO and existing MIMO configurations is presented in Table 3, where isolation, ECC, and DG value is compared. The intended MIMO antenna shows near the ideal value of both parameters compared to other existing MIMO antennas. This excellent feature makes the intended MIMO antenna a compatible candidate for mm-wave 5G applications.

Table 3 Comparison of intended MIMO antenna and existing MIMO antenna.
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Conclusion

An HRIMDA for mm-wave 5G applications is exhibited in this paper. A horizontal H-shape (HRIM) unit cell is modelled, and the unit cell’s metamaterial property is utilized for gain enhancement. The designed antenna is fabricated and verified by measurement, where the single fabricated antenna has a small size of 0.77 λ0 × 1.38 λ0 × 0.0195 λ0. The proposed HRIMDA shows − 10 dB (49.2 BW%) impedance bandwidth from 23 to 38 GHz frequency. The antenna gain notably improves by incorporating HRIM in the typical antenna and maximum enhancement by 3.5 dBi. The MIMO performance of the proposed HRIMDA is also investigated, which shows near the ideal value of ECC (< 1 × 10–3) and DG (> 9.9). Therefore, the proposed HRIM dipole antenna’s excellent features make it an excellent candidate for upcoming advanced 5G applications, such as smart cities, autonomous vehicles, telemedicine, and enhanced mobile broadband. These applications require high data rates, low latency, and reliable connectivity, all of which are supported by the high gain and wide bandwidth of the HRIMDA. Additionally, the compact size and simple design of the antenna make it suitable for integration into various portable and fixed wireless devices. Finally, the HRIMDA represents a significant advancement in antenna technology for 5G mm-wave applications, combining high performance with practical design considerations. Its development paves the way for more efficient and reliable wireless communication systems, meeting the ever-growing demand for high-speed connectivity in our increasingly connected world.