Introduction

With the continuous advancement of communication technologies, circularly polarized (CP) antennas have been widely adopted in satellite communications, mobile/wireless networks, and millimeter-wave systems due to their polarization compatibility, multipath interference robustness, and signal transmission stability1,2,3,4,5,6. To meet the growing demand for high-capacity, high-speed data links in satellite communications, the development of high-gain, broadband CP array antennas has become a significant research focus in both academic and industrial sectors.

In the research evolution of broadband high-gain CP antennas, microstrip antennas7,8,9,10,11,12,13,14, slot antennas15,16,17,18,19,20, cavity-backed antennas21,22,23,24,25,26, and magneto-electric dipole antennas27,28,29,30 have constituted the conventional mainstream approaches. A sequentially rotated (SR) dual-layer microstrip antenna array was demonstrated in7, achieving an axial ratio bandwidth of 63.7%. Ref. 15 introduced a 1\(\times\)8 dual-CP antenna array that attained peak gains of 17.3 dBic and 17.9 dBic under left-handed (LHCP) and right-handed circular polarization (RHCP) operation, respectively. Ref. 21 designed a \(1 \times 3\) cavity-backed antenna array utilizing magneto-electric dipoles, which achieves broadband high-gain circularly polarized radiation while maintaining low sidelobe levels. However, the cross-polarization discrimination (XPD) remains below −18 dB, reflecting strong mutual coupling between elements that consequently restricts the polarization purity and overall array performance.

The Substrate Integrated Cavity (SIC) antenna31,32,33,34,35,36,37 represents an implementation form of cavity-backed antennas, which constructs a waveguide cavity structure within the dielectric substrate through metallized vias. This type of antenna offers advantages such as effective surface wave suppression, high gain, and low profile. However, when multiple SIC antenna units are configured into an array with small element spacing, surface waves can still be excited and propagate along the dielectric substrate. When the phase velocity of the surface wave approaches that of the quasi-TEM wave, the electromagnetic coupling between adjacent elements is intensified, leading to degradation in isolation and XPD performance.

Ref. 31 reported a \(4 \times 4\) SIC array achieving an impedance bandwidth of 31% and a peak gain of 21 dBic, demonstrating excellent broadband high-gain CP radiation characteristics. However, the 3-dB axial ratio (AR) bandwidth of this design was only 24%. According to Ref. 32, an \(8 \times 8\) SIC array realized CP radiation through two parasitic strip patches tilted at \(45^\circ\), achieving a peak gain of 26.1 dBic. Nevertheless, its usable bandwidth remained limited at 21.72%, which is still insufficient. Ref. 33 proposed a \(2 \times 2\) SIC CP antenna array that achieved a usable overlapping bandwidth of 35.4% using only two dielectric substrate layers, exhibiting outstanding integration and broadband performance.Despite these achievements, the XPD values reported in Refs. 31,32,33 were below −15 dB, −18 dB, and − 20 dB, respectively, indirectly reflecting strong inter-element coupling. Although these values meet basic application requirements, there remains significant potential for further improvement to achieve purer CP radiation characteristics.

This paper presents a Ka-band broadband high-gain circularly polarized antenna array with high cross-polarization discrimination. Initially, a Ka-band high-gain linearly polarized element loaded with an array decoupling surface (ADS) is designed. By introducing magnetoelectric dipole parasitic antennas on the SIC radiating aperture and exciting the higher-order TM211 mode within the SIC, high-gain linear polarization radiation is achieved. Subsequently, a double-layer substrate integrated waveguide (SIW) sequential phase-shift feeding network is designed, which generates a 90° phase delay by varying the lengths of the four feeding paths. The proposed design innovatively adopts a four-layer stacked dielectric substrate structure and achieves breakthroughs in wide impedance bandwidth, high isolation, and high polarization purity through the co-design of a sequentially rotated magneto-electric dipole array and an ADS. The main contributions of this work are summarized as follows:

  • A novel high-integration stacked configuration: the proposed design innovatively employs a four-layer stacked substrate structure that incorporates SIC magneto-electric dipole elements in a 2×2 SR layout. This architecture achieves a breakthrough in balancing structural compactness and radiation performance, enabling a highly integrated solution for space-constrained millimeter-wave applications.

  • Exceptional broadband circular polarization performance: the antenna demonstrates groundbreaking broadband characteristics, with both simulated and measured \(-10\) dB impedance bandwidth exceeding 30%. Remarkably, the 3 dB axial ratio bandwidth fully encompasses the impedance bandwidth, ensuring stable circularly polarized radiation across the entire operational spectrum.

  • Effective mutual coupling suppression: through the integration of an ADS, the design achieves remarkable inter-element isolation improvement of approximately 10 dB, reaching an exceptional XPD ratio better than \(-25\) dB throughout the operating band. This innovation effectively addressing the longstanding challenge of mutual coupling in dense millimeter-wave arrays, thereby significantly advancing the state-of-the-art in polarization purity.

  • Superior gain stability and system compatibility: the array exhibits outstanding gain stability with a peak gain of 12.47 dBic and minimal fluctuation less than 3 dB within the 25.95–33.14 GHz range (24.34% gain bandwidth). This combination of wide bandwidth, high polarization purity, and exceptional gain stability makes the array particularly suitable for 5G millimeter-wave communication systems requiring reliable high-speed data links.

Fig. 1
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3-D exploded view of the antenna element.

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Antenna element design

Element configuration

Figure 1 shows the structural model of the Ka-band linearly polarized antenna element, which comprises three layers: the top ADS layer, the middle substrate integrated cavity (SIC) radiating layer, and the bottom substrate integrated waveguide (SIW) feeding layer. Layer 1 employs Rogers 4003 (\(\varepsilon _r = 3.55\), \(\tan \delta = 0.0027\)), whose relatively high dielectric constant contributes to reducing the ADS structure size and improving integration density. Layers 2 and 3 utilize Rogers 5880 (\(\varepsilon _r = 2.2\), \(\tan \delta = 0.0009\)), whose lower dielectric constant helps suppress surface wave propagation and reduce mutual coupling, while the extremely low loss tangent makes it particularly suitable for millimeter-wave applications, effectively minimizing feeding network loss and thereby ensuring antenna gain and efficiency.

The substrate of Layer 1 is etched with a \(2\times 2\) array of segmented multilayer square-loop decoupling structures. Through precise optimization of the loop geometry, dimensional parameters, and substrate thickness, these structures generate reflected waves with specific amplitude and phase characteristics at the array ports to effectively cancel the coupled waves between elements. The metallic square loops are divided into discontinuous sub-elements to minimize the shadowing effect on the radiation pattern of the underlying antenna caused by the electrically large decoupling surface. The substrate of Layer 2 is loaded with four larger metallized vias, four rows of smaller metallized vias, and two symmetrically positioned metal rectangular patches. These metal patches, the larger vias, and the central gap between them collectively form a magnetoelectric dipole antenna, while the four rows of smaller metallized vias constitute the SIC. Layer 3 incorporates an SIW feeding network composed of a bottom metal ground plane, a Rogers 5880 dielectric substrate with three rows of metallized vias, and a top metal plate etched with feeding slots. More detailed structural parameters are provided in Table 1.

Table 1 Antenna structural parameters.
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Fig. 2
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Magnetoelectric dipole surface current distribution.

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Fig. 3
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Effect of parameter adjustment on \(S_{11}\).

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Working principle of the element

For millimeter-wave antenna arrays, the excessively small element spacing causes post-fabrication integration of ADS to significantly impact bandwidth and radiation patterns while complicating tuning procedures, therefore the ADS is synchronously designed during the element design stage. The proposed methodology first optimizes the S-parameters and radiation patterns of unit cells loaded with the ADS. These optimized cells are then incorporated into the full array to verify the ADS’s efficacy in mutual coupling suppression while ensuring compatibility with the overall system performance.

Fig.2 illustrates the current distribution characteristics of the magnetoelectric dipole antenna operating at the center frequency of 28 GHz. As observed, the antenna currents are predominantly concentrated along the edge regions of the two rectangular metallic patches and within the interstitial gap between them. The strongest current density occurs on both sides of the central gap, exhibiting equal-amplitude anti-phase distribution characteristics that show excellent agreement with the classical half-wavelength dipole antenna current pattern. The currents on the two rectangular metallic patches maintain co-directional flow with amplitude tapering from the adjacent edges toward the distal ends, fully consistent with the theoretical prediction of dipole-pair formation through gap-coupled feeding. Notably, the substantially weaker current magnitude on the patches compared to the gap edges, combined with their orthogonal relationship, satisfies the fundamental condition for magnetoelectric dipole operation and enables complementary radiation patterns.

The operating frequency band of the antenna element is governed by the length of the transverse slot, the dimensions of the rectangular radiating patch, and the geometric parameters of the four large metalized vias. As illustrated in Fig. 2, the current distribution along the edges of the rectangular patch forms an electric dipole radiation mechanism. Its fundamental resonant frequency \(f_e\) is primarily determined by the patch length \(w_{p1}\). In the design process, \(w_{p1}\) is first optimized to bring \(f_e\) close to the target center frequency of 28 GHz, thereby establishing a stable electric dipole radiation mode.The four large metalized vias, together with the central gap, form a closed current loop that constitutes the physical structure of a magnetic dipole. The via radius \(R_1\) directly determines the equivalent inductance of the metallic posts. Based on the inverse relationship between inductance and radius, the inductance decreases as \(R_1\) increases. The via spacing \(d_3\) controls the area enclosed by the current loop, which is proportional to the square of the spacing. According to the magnetic dipole moment theory \(m = I \cdot A\), increasing \(d_3\) can significantly enhance the radiation strength of the magnetic dipole but simultaneously reduces its resonant frequency.

Based on the above operational principles, the magnetic dipole’s resonant frequency \(f_m\) is determined collectively by the equivalent inductance and capacitance of the loop. A decrease in \(R_1\) increases the inductive reactance of the posts, forcing \(f_m\) downward. Conversely, an increase in \(d_3\) further reduces \(f_m\) by decreasing the loop capacitance. Achieving a broad bandwidth requires the precise control of the magnetic dipole resonance frequency \(f_m\) via the co-optimization of parameters \(R_1\) and \(d_3\). This tuning aims to bring the magnetic dipole resonance \(f_m\) close to the electric dipole resonance \(f_e\) while maintaining a suitable separation, ultimately achieving a double-resonant peak structure. The impact of these parameter adjustments on the impedance matching performance is demonstrated in Fig. 3.

Fig. 4
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Antenna element simulation results.

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Fig. 5
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The simplified \(2 \times 2\) array configuration without an integrated feeding network.

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Fig. 6
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Surface current distribution on the array antenna without ADS at 27.735 GHz.

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Fig. 7
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Surface current distribution on the array antenna with ADS at 27.735 GHz.

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Fig. 8
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Comparative analysis of the isolation curves for the antenna array w/ and w/o ADS.

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Simulated results

Figure 4a presents the final optimized simulation results of the antenna element, which exhibits two distinct resonant points at 24.69 GHz and 31.22 GHz, demonstrating excellent broadband radiation characteristics with a reflection coefficient below −10 dB across 23.15–32.65 GHz and an impedance bandwidth of 34.53%.The peak gain reaches 8.63 dBi, while maintaining gain levels above 8 dBi throughout 25.91–33.84 GHz. Comparative analysis of ADS-removed configurations reveals that although the ADS structure slightly degrades impedance matching, it significantly enhances antenna gain within the operational band. Figure 4b presents the 2D radiation patterns, which exhibit an asymmetric steep depression at the bottom. This phenomenon originates from electromagnetic blockage and scattering by the ADS layer but becomes negligible after the array is fully integrated. It is noteworthy that during the unit cell design phase, the ADS layer primarily serves to extend the operational bandwidth and enhance radiation gain, while its decoupling functionality in array configurations will be comprehensively examined in subsequent research.

ADS decoupling performance

To evaluate the effectiveness of the ADS in suppressing mutual coupling between antenna elements and enhancing cross-polarization performance, the magneto-electric dipole antenna elements were arranged in a SR configuration to form a \(2 \times 2\) array architecture, as illustrated in Fig. 5. Experiments were conducted on a simplified array structure without the integrated feeding network to specifically isolate and analyze the impact of the ADS on electromagnetic field distribution.Figs. 6 and 7 illustrate the surface current distributions on the antenna radiating layer at 0°, 90°, 180°, and 270°with and without ADS integrated. Without the ADS (Fig. 6), strong coupling currents are observed in the regions adjacent to the elements, accompanied by disordered current vector directions, indicating significant mutual coupling. This chaotic current distribution degrades the antenna’s polarization purity. Upon integrating the ADS (Fig. 7), the coupling currents in the inter-element regions are effectively suppressed, resulting in a more ordered current distribution. This demonstrates the significant improvement in mutual coupling characteristics achieved by the ADS. Simultaneously, a distinct counterclockwise rotation of the current direction along the normal axis is clearly observed, verifying the mechanism of RHCP radiation.

The simulation results in Fig. 8 demonstrate that incorporating the ADS enhances the inter-element isolation by approximately 10 dB, achieving an overall isolation level of \(-32\) dB. This improvement not only reduces energy coupling loss but, more importantly, restores the symmetry of the electromagnetic field distribution by suppressing mutual coupling effects, thereby fundamentally enhancing the antenna’s cross-polarization purity. The radiation patterns in Fig.9 demonstrate that the cross-polarization component is significantly suppressed in the array with the integrated ADS, compared to the configuration without ADS.

Wideband CP antenna array

2\(\times\)2 wideband CP antenna array design

Building on the unit design and ADS effectiveness validation, a 2\(\times\)2 SR fed SIC array antenna is developed to realize broadband CP performance. As shown in Fig. 10, the array employs a four-layer dielectric substrate configuration: the top layer comprises a 4\(\times\)4 ADS surface; the second layer integrates a 2\(\times\)2 magnetoelectric dipole array arranged in sequential rotation (wherein the window-type SIC from the unit design is optimized into orthogonal metallized via isolation strips for array integration); and the third and fourth layers together form a dual-layer SR feeding network. This network is centrally fed through a coaxial-to-SIW transition, with energy coupling between the dual-layer feeding networks achieved via transverse slots. The four-layer substrate possesses thicknesses of 1.1 mm, 0.78 mm, 0.78 mm, and 0.762 mm, respectively. The overall dimension of the antenna array is 22 mm \(\times\) 22 mm \(\times\) 3.422 mm, corresponding to an electrical size of \(2.03\lambda _0 \times 2.03\lambda _0 \times 0.32\lambda _0\) at the operating frequency.

Fig. 9
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Comparative analysis of radiation patterns for the antenna array with and without ADS.

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Fig. 10
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3-D exploded view of the 2×2 antenna array.

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Fig. 11
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Double-layer SIW feeding network.

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Figure 11a–c details the feeding network structure of the dual-layer SIW configuration, which employs a progressive path difference distribution scheme to achieve broadband circularly polarized excitation. Specifically, the upper-layer feeding network incorporates an incremental length difference of 1.75 mm along the counterclockwise direction for each feeding path (Fig.11a), while the lower-layer network is engineered with a compensatory path differential of 0.87 mm (Fig.11b). This hierarchical differential architecture operates on the transmission phase accumulation principle, whereby precise control over the electromagnetic wave propagation path length within the SIW medium generates a nominal 90° phase lag between adjacent output ports at the center frequency of 27.735 GHz, thereby fulfilling the core requirement for phase consistency in sequentially rotated arrays.As demonstrated by the measured results in Fig. 11d, the feeding network maintains reflection coefficients below \(-10\) dB across the 23–32 GHz band with phase differences of 90\(^\circ\) (± 10\(^\circ\)) between ports, fully meeting the requirements for broadband circularly polarized arrays.

Fig. 12
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Photograph of the proposed antenna array.

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Fig. 13
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Simulated and measured radiation patterns of the antenna array.

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Fig. 14
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Antenna array simulation and measured results.

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Measured results

To validate the performance of the proposed broadband CP array, the antenna was modeled and simulated using ANSYS HFSS, and a prototype of the array was fabricated for experimental measurement and analysis.The prototype of the array antenna is illustrated in Fig. 12, where four substrate layers are integrated and reinforced using eight nylon screws. Fig.13 shows the simulated and measured normalized radiation patterns in both the xoz- and yoz-planes at 24 GHz, 28 GHz, and 32 GHz, from which reasonable agreement can be observed.

Figure 14a presents the simulated and measured results of the S\(_{11}\) and AR for the antenna array. The simulated results indicate an impedance bandwidth of 23.45–32.02 GHz (30.89% relative bandwidth) at the −10 dB level, while the measured impedance bandwidth spans 23.46–31.85 GHz (30.33% relative bandwidth). In terms of axial ratio performance, the simulated 3-dB AR bandwidth covers 22.35–32.41 GHz (36.74% relative bandwidth), and the measured 3-dB AR bandwidth extends from 22.48 GHz to 32.19 GHz (35.52% relative bandwidth). The overlapping bandwidth between the AR and impedance performance is 30.89% for simulations and 30.33% for measurements.

Figure 14b presents the simulated and measured gain and radiation efficiency of the proposed antenna. The simulated results indicate a peak gain of 12.54 dBic, with a gain variation below 3 dB across 27.25–33.27 GHz, corresponding to a 19.89% gain bandwidth. Moreover, the XPD exceeds −25 dB over 24.07–33.87 GHz. Measurement results show a peak gain of 12.47 dBic with gain fluctuations below 3 dB throughout 25.95–33.14 GHz, yielding a 24.34% gain bandwidth, while the XPD remains better than −25 dB within 23.19–33.17 GHz. The antenna exhibits high radiation efficiencies, with the simulated and measured results achieving 75% and 70%, respectively. The slight discrepancies between simulated and measured results are mainly attributed to fabrication tolerances of the dielectric substrates, variations in material parameters, and alignment inaccuracies introduced by the 2.92-KHD feed connector.

Comparison and discussion

Table 2 summarizes the performance comparison between the proposed broadband CP antenna array and existing designs in chronological order of publication, highlighting its comprehensive advantages. Among the solutions employing the 2×2 SIC array configuration, Ref. 33,38,39 are representative and are compared in detail. Ref. 38 utilizes LTCC technology and radiates via an E-shaped planar inverted-F antenna, where the constructed \(2 \times 2\) subarray employs the SIC to excite higher-order modes and a coupling disk to perturb the field distribution, thereby automatically generating a 90° phase difference with equal-amplitude feeding for self-sequential rotation.This design achieves a high aperture efficiency of 71% and a compact size of 2.03\(\lambda _0\) × 2.03\(\lambda _0\) × 0.32\(\lambda _0\) within the 110–170 GHz band, making it suitable for terahertz communication systems. Ref. 39 proposes an antenna array with a single-layer structure having a low profile of only 0.035\(\lambda _0\); by integrating a circular patch resonator within the SIC, it achieves stable radiation characteristics and integrated filtering functionality, with an aperture efficiency exceeding 90% and isolation better than −25 dB. However, the impedance bandwidths of these designs are less than 20%, specifically 10.4% for Ref. 38 and 5.94% for Ref. 39. In contrast, the proposed \(2 \times 2\) SIC array antenna employs a four-layer stacked substrate and a novel sequential rotation feeding network, achieving an impedance bandwidth of 30.33% (23.46–31.85 GHz) and an AR bandwidth of 35.52% (22.48–32.19 GHz). Although the \(2 \times 2\) SIC array in Ref. 33 achieves a wide overlapping bandwidth of 35.4% (27.92–39.93 GHz) through multi-mode resonance, its measured peak gain is relatively low at 11.7 dBic.

More importantly, the XPD performance of all compared designs is inferior to the proposed one. By incorporating an ADS, the proposed design effectively suppresses mutual coupling between elements, resulting in an XPD better than −25 dB across the entire operating band. Simultaneously, the peak gain of 12.47 dBic and the usable bandwidth of 30.33% demonstrate excellent performance among similar circularly polarized array designs, effectively meeting the requirements of 5G communication systems for wide bandwidth and high polarization purity.

Table 2 Performance comparison of previous CP antenna arrays.
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Conclusions

This paper successfully demonstrates a high-performance broadband CP antenna array operating in the Ka-band. The core innovation is embodied in a co-design approach that integrates a \(2 \times 2\) SR SIC magneto-electric dipole array with a top-layer ADS through a four-layer stacked architecture, achieving a breakthrough in both radiation performance and integration level. Specifically, a novel double-layer SIW SR feeding network is innovatively designed to realize wideband and precise 90-degree phase delay. The incorporation of the ADS structure enhances the XPD to better than -25 dB, a significant improvement over conventional designs. By virtue of its highly compact physical profile, exceptional wideband characteristics, and outstanding polarization purity, this antenna emerges as an ideal solution for high-speed point-to-point backhaul links in 5G millimeter-wave communication systems and satellite communication terminals. It is particularly noteworthy that the underlying ADS design philosophy offers a critical reference for developing future highly integrated millimeter-wave antenna systems.