Introduction

Monopulse antennas are predominantly used in tracking radar systems for a variety of applications, including air traffic control, cosmic debris detection, and satellite tracking. This is due to their ability to accurately and efficiently determine the direction and location of a target. In a practical monopulse tracking system, the signals received from the sum and difference channels allow the system to gather information about a target’s position. This requires a radiation pattern with suitable peak and sharp null.

There are two main approaches to the design of monopulse feeds. The most straightforward conventional monopulse tracking feed system includes a feed with four or more antennas and a comparator network1. A monopulse comparator network is employed to combine the received signals from the antennas to generate the sum and difference radiation patterns. An alternative approach is to use a multimode horn antenna2. The main disadvantage of this approach is the narrow bandwidth of the monopulse performance (up to 3%)3,4,5,6. For example, a multimode corrugated horn feed with a monopulse performance operating at 35 GHz and a bandwidth of 2.3% was presented in5.

To create a compact monopulse antenna, open ended waveguide and horn antennas with small apertures can be placed closely because of their small size. However, the main disadvantages of these structures are limited bandwidth, single polarization and non-symmetric beam in the principal planes7,8,9. In recent years, some efforts have been made to design a wideband monopulse feed10,11,12,13. For instance, an array of Vivaldi antennas has been used in10 to achieve monopulse performance over the frequency ranges of 3.1–10.6 GHz. In12, an ultrawideband compact monopulse feed has been proposed, covering 5.5–9 GHz band. The main drawback of these antennas is their single polarization.

In certain practical radar applications, dual-polarized or 45° slant-polarized monopulse antennas are required due to their immunity to interference. In recent years, various types of dual-polarized monopulse antennas have been developed14,15,16,17,18,19,20,21,22 for use in the scenarios in which the target can change the polarization of the wave or when Faraday rotation occurs. Such tracking capabilities are particularly useful in satellite tracking, mobile satellite communications and very small aperture terminals. In15, a dual-polarized monopulse array with highly self-compact configuration has been proposed with a bandwidth of 12.5% in the Ku-band. A dual-polarized 2-D monopulse conical conformal antenna has been reported in18 with an impedance bandwidth of 22.2% from 8.96 to 11.2 GHz, a sum-beam gain of 9–11 dBi, a difference-beam gain of 5.8–7.7 dBi, and a null-depth of around − 25 dB. A dual-polarized monopulse feed excited by a gap waveguide-based comparator has been reported in20. The measured results showed that the developed feed operating at 34 GHz featured 20.9% fractional bandwidth and the null depth of the difference beam of -30 dB.

The 8–18 GHz frequency band, encompassing the X-band and Ku-band, is crucial for various advanced applications due to its unique properties. This band is widely used in tracking systems, precise target detection, weather monitoring, air traffic control, and defense applications. The ability of these frequencies to penetrate through various atmospheric conditions while maintaining high data rates makes the 8–18 GHz band indispensable in modern communication, sensing technologies and tracking radars.

A comprehensive review of the literature reveals a significant lack of information on ultrawideband dual- or slant-polarized monopulse feeds for the 8–18 GHz frequency band. To the best of the authors’ knowledge, none of the monopulse antennas reported in previous studies meet the requirements for both slant polarization and ultrawideband performance simultaneously. This work aims to address this gap by developing an innovative ultrawideband, compact, and slant-polarized monopulse feed using wideband slant-polarized radiators in a 2 × 2-element configuration. The novelty of the proposed monopulse feed design lies in the use of four ultrawideband end-launched diagonal horn antennas. The diagonal horn antennas offer the advantage of providing slant polarization while maintaining an acceptable sidelobe level (SLL) and a symmetrical beam in the 2 × 2 array. By introducing an ultrawideband transition between the rectangular waveguide and coaxial line, the proposed diagonal horn antenna achieves excellent impedance matching over the 8 to 18 GHz frequency band. These features make the proposed solution particularly suitable for ultrawideband tracking radars.

Single-element antenna design

A diagonal horn antenna

The main building block in the proposed ultrawideband monopulse feed is a diagonal horn antenna. The structure of a diagonal horn antenna and its detailed dimensions are depicted in Fig. 1(a). The diagonal horn antenna is a specialized type of microwave horn antenna, notable for its unique design where the output aperture is angled 45° from the feed waveguide. This type of horn antenna usually features a square aperture, promoting a more consistent phase distribution across the aperture, which reduces phase errors and increases directivity and gain. The diagonal horn antenna was first introduced in23 and then employed for various microwave and mmWave applications24,25,26. The diagonal horn antenna can support two orthogonal TE10 and TE01 modes to create almost rotational patterns for smaller values of elevation angle. The far-fields of the diagonal horn in H- and E-planes are given as follows:

$${\rm H-plane}:\:{E}_{\theta\:}=0,\:\:\:\:\:\:,{E}_{\phi\:}=-\frac{jk{e}^{-jkr}}{4\pi\:r}\sqrt{2}\:\left(1+\text{cos}\theta\:\right)$$
(1)
$${\rm H-plane}\:{E}_{\phi\:}=0\:\:\:\:\:\:\:\:\:\:{E}_{\theta\:}=\frac{jk{e}^{-jkr}}{4\pi\:r}\sqrt{2}\:\left(1+\text{cos}\theta\:\right)$$
(2)

where k is the propagation constant in the free space. The diagonal horn antenna has drawn significant interest as a high-efficiency primary radiator in satellite communication and imaging applications. This is due to its symmetric beam, low side lobe level (SLL), and low cross-polarization level.

The rotation of the electric field distribution within the flare is a key operational aspect of diagonal horn antenna. As illustrated in Fig. 1(b), the flare section induces the electric field to rotate as it propagates through it and the TE10 mode in the feed waveguide is transformed into a combination of two orthogonal TE10 and TE01 modes in the square aperture of the antenna. This feature allows the desired 45° slant-polarized radiation pattern to be the achieved simply by a diagonal horn antenna with square aperture. The operating mechanism is similar to that of waveguide twists. However, the bandwidth of a conventional waveguide twist is limited to about 40%27, while it is desirable to be able to cover the band extending from 8 to 18 GHz, corresponding to a 76.9% fractional bandwidth.

Fig. 1
figure 1

(a) Configurations of a waveguide-fed diagonal horn antenna. (b) Electric field distribution in different cross-sections of the antenna from feed waveguide to square aperture of antenna.

Full size image

B ultrawideband transition

In order to excite the diagonal horn antenna with a standard SMA connector, an ultrawideband transition between the coaxial line and the rectangular waveguide is required. Figure 2(a) illustrates the proposed transition structure, which extends the rectangular waveguide to align with the SMA connector. For practical use in a monopulse system, it is preferable to place the connectors on the back wall of the waveguide; hence, the transition is implemented in an end-launched form. In this structure, to achieve wide bandwidth, a stepped ridge is inserted on the bottom wall of the waveguide. Additionally, a rectangular groove with dimensions of Lg × Wg × Hg is created in the back wall of the waveguide around the connector junction. Designing the waveguide to coaxial line transition typically involves calculations based on mode conversion and impedance matching theories28,29,30. The characteristic impedance of a double-ridge waveguide is given by.

Fig. 2
figure 2

(a) Configurations of an ultrawideband coaxial to waveguide transition. (b) Cut view of end-launched diagonal horn antenna.

Full size image
$${Z_0}\,=\,\frac{{Z_{0}^{\infty }}}{{\sqrt {1 - {{\left( {\frac{\lambda }{{{\lambda _c}}}} \right)}^2}} }}$$
(3)

where λc is the cutoff wavelength and Z0∞ is the characteristic impedance at infinity is given by

$$Z_{0}^{\infty }\,=\,\frac{{\pi {\eta _0}{H_1}}}{{{\lambda _c}\left( {{X_1}+{X_2}} \right)}}$$
(4)
$${X_1}=\,\frac{{2{H_1}}}{{{\lambda _c}}}{\cos ^2}\left( {\frac{{\pi {W_w}}}{{{\lambda _c}}}} \right)\ln \csc \left( {\frac{{\pi {H_1}}}{{2{H_w}}}} \right)+\frac{{\pi {W_r}}}{{2{\lambda _c}}}+\frac{1}{4}\sin \left( {\frac{{2\pi {W_r}}}{{{\lambda _c}}}} \right)$$
(5)
$${X_2}\,=\,\frac{{{H_1}}}{{{H_w}}}\frac{{{{\cos }^2}\left( {\frac{{\pi {W_r}}}{{{\lambda _c}}}} \right)}}{{\sin \left( {\frac{{\pi ({W_w} - {W_r})}}{{{\lambda _c}}}} \right)}}\left( {\frac{{\pi ({W_w} - {W_r})}}{{2{\lambda _c}}} - \frac{1}{4}\sin \left( {\frac{{\pi ({W_w} - {W_r})}}{{{\lambda _c}}}} \right)} \right)$$
(6)
Fig. 3
figure 3

Variation of impedance of a double ridge waveguide for different values of ridge spacing.

Full size image
Fig. 4
figure 4

(a) S-parameters and (b) group delay of the proposed transition in a back-to-back configuration.

Full size image

Based on (3) to (6), Fig. 3 shows the variation of impedance of the double ridge waveguide for different values of ridges spacing. According to the results, utilizing multi-stepped ridged sections is an efficient approach to achieve the transition from an impedance of 50 Ω to a few-hundred impedance. These ridged sections can be arranged to maintain a consistent reflection across the frequency range of 8 to 18 GHz.

To properly design the transition, a back-to-back structure is simulated and optimized. The optimization of geometrical parameters is essential to convert the rectangular waveguide mode into the coaxial line mode effectively in the wide frequency band of interest. This optimization ensures proper matching at the input port and minimizes insertion loss across the desired frequency range. In the simulation model in CST, Aluminum with an electrical conductivity of 3.6 × 107 S/m is used. The simulation results for the S-parameters and group delay of the optimized transition in a back-to-back configuration are shown in Fig. 4. The input reflection coefficient of the structure is less than − 15 dB from 8 to 18 GHz. It is evident that the group delay is almost constant throughout the operating frequency band. This results in the signal experiencing the same delay at all frequencies, thereby reducing distortion and preserving the signal’s integrity.

C complete single-element antenna

The configuration of end-launched diagonal horn antenna is illustrated in Fig. 2 (b). The optimized values for its geometrical parameters are given in Table 1. The simulated |S11| and realized gains of the complete end-launched diagonal horn antenna versus frequency are illustrated in Fig. 5. The antenna exhibits 76.9% fractional bandwidth from 8 to 18 GHz for |S11| < -13 dB with a peak realized gain of 12.82 dBi at 18 GHz. The radiation patterns at 8, 13, and 18 GHz in ϕ = 0° or ϕ = 90° planes for both co- and cross-polarized components are displayed in Fig. 6. Notably, the radiation patterns are symmetric and exhibit half-power beamwidth of 41-77° in both principal planes, and the cross-polarization discrimination in the boresight is higher than 50 dB.

Table 1 Optimized values for design parameters of end-launched diagonal horn antenna.
Full size table
Fig. 5
figure 5

Simulated |S11| and realized gain of end-launched diagonal horn antenna.

Full size image
Fig. 6
figure 6

Radiation patterns of diagonal horn antenna ϕ = 0° (or ϕ = 90°) at different frequencies (a) Co-polarized and (b) cross-polarized components.

Full size image

Monopulse Feed design

The primary goal of the present work is to use of a 2 × 2-element configuration using four horn antennas to achieve ultrawideband monopulse feed with slant 45 polarization. Initially, the simplest option appears to be using four ultrawideband double-ridge horn antennas, each rotated by 45 degrees, as illustrated in Fig. 7(a). In this intuitive and simple design ensuring broadband operation is straightforward, as various ultrawideband horn antennas with excellent bandwidth and radiation characteristics have been reported in the literature31,32,33. This approach has some primary challenges. Firstly, the rotation required for slant polarization results in asymmetric radiation patterns for each double-ridge horn antenna in the vertical and horizontal planes. Secondly, most ultrawideband horn antennas have large apertures and this rotation increases the spacing between the radiation centers of adjacent elements in the array, worsening the SLL of the monopulse antenna and potentially leading to grating lobes. An alternative option is to use of diagonal quad-ridge horn antennas discussed in34] and [35. There are three key points to consider with this approach. Firstly, these antennas unlike conventional diagonal horns (without ridges in the flare section) do not have exact identical radiation patterns and in the E- and H-planes. Secondly, the aperture dimensions of a diagonal quad-ridged horn antenna with a gain of approximately 8–13 dBi over 8–18 GHz band are 35 mm × 35 mm, which is approximately 1.6 times the aperture dimensions of the proposed diagonal horn (22 mm × 22 mm). This results in increasing the spacing between the radiation centers of adjacent elements in the array and worsening the SLL of the monopulse feed. Thirdly, from the practical point of view, using diagonal quad-ridge horn antennas tangentially and side by side in a monopulse feed poses a significant challenge. Unlike the proposed end-launched antenna, the feed ports of diagonal quad-ridge horn antennas are located at the corners, leaving insufficient space for connecting the feed connectors.

Fig. 7
figure 7

Configuration of 2 × 2-element arrays of (a) double-ridged, (b) diagonal quad-ridged and (c) diagonal horn antennas (All three figures use the same scale).

Full size image

To illustrate the SLL challenge in array configuration, note that the aperture dimensions of a double-ridged horn antenna with a gain of approximately 8–13 dBi are 40 mm × 28 mm31. As shown in Fig. 7(a) and 7(b), arranging four 45° rotated double-ridged pyramidal horn or diagonal quad-ridged horn antennas in a 2 × 2 array results in a minimum center-to-center spacing of S1 = 40 mm S2 = 35 mm, respectively. In these cases, the SLL of arrays is significantly increased. To address this issue, we propose a new design in which four diagonal horn antennas are used instead of double-ridged pyramidal or diagonal quad-ridged horn antennas, as illustrated in Fig. 7(c). This setup not only reduces the center-to-center spacing by almost half, but also results in a symmetrical beam (S3 = 22 mm). To demonstrate the superiority of the proposed new design, Fig. 8 compares the SLL of 2 × 2-element arrays using double-ridged, diagonal quad-ridged and the proposed diagonal horn antennas. It is evident that utilizing an array of diagonal horn antennas significantly improves the SLL.

Fig. 8
figure 8

Comparison of SLLs of 2 × 2-element arrays of double-ridged, diagonal quad-ridged and diagonal horn antennas.

Full size image

The proposed ultrawideband 45° slant-polarized monopulse feed, as illustrated in Fig. 9, consisting of four end-launched diagonal horn antennas to address the aforementioned challenges related to SSL, beam symmetry and space for placing the feed connectors. Each horn is fed from the backside using a multi-step ultrawideband double- ridge rectangular waveguide adaptor and a standard SMA connector. The radiating apertures are excited by 45° electric fields to achieve slant polarization. The signal from the output ports of the antennas enters a commercial circuit-based comparator network. The frequency band of interest for the feed performance is 8–18 GHz, corresponding to a 76.9% fractional bandwidth.

Fig. 9
figure 9

Configuration of proposed ultrawideband 45° linearly polarized monopulse feed in perspective and front views.

Full size image

Fabrication and measurement

A measurement results

To validate the proposed design, a prototype of the ultrawideband monopulse feed was fabricated using low-loss aluminum. Figure 10(a) shows photographs of the fabricated prototype. As previously mentioned, although the feed is designed to operate from 8 to 18 GHz, conventional hollow waveguide-based monopulse comparators could not be used in the measurements because they cannot cover the entire feed’s bandwidth. Consequently, commercial circuit-based delta-sigma hybrids (MCD020180H6M) with the operating frequency band from 2 to 18 GHz were employed in the measurements, as depicted in Fig. 10(b).

Fig. 10
figure 10

Photographs of fabricated monopulse feed. (a) assembled structure. (b) Antenna under test.

Full size image

Far-field radiation pattern measurements were conducted in a high-frequency anechoic chamber, as shown in Fig. 10(b). The simulated and measured |S11| and realized gain of the slant-polarized monopulse feed in sum and difference modes in the frequency range of 8–18 GHz are compared in Fig. 11. The measured |S11| is below -11.27 dB over the desired bandwidth. The measurements indicate that the measured realized gain in the sum mode is between 12.33 and 17.92 dBi which is approximately 3 dB higher than the gain in the difference mode. A reasonable agreement is observed between the simulated and measured results, with some negligible discrepancies possibly due to losses in connectors, cables, circuit-based delta-sigma hybrids, and fabrication and measurement setup errors. The simulated and measured normalized radiation patterns in the sum and difference modes at 8, 13 and 18 GHz are plotted in Fig. 12. The measured amplitude imbalance and null-depth of difference-pattern are less than 0.26 and -32 dB, respectively. Some variations can be seen between the simulation and measured plots in the back lobes in some frequencies, attributed to the errors of measurement setup in the anechoic chamber.

Fig. 11
figure 11

Simulated (solid) and measured (dashed) |S11| and realized gain for ultrawideband monopulse feed antenna.

Full size image
Fig. 12
figure 12

Simulated and measured normalized radiation patterns of monopulse feed in sum and difference modes at different frequencies. (a) and (b) 8 GHz. (c) and (d) 13 GHz. (e) and (f) 18 GHz.

Full size image

Discussion

To highlight the superiority of the proposed monopulse feed, its performance is compared with similar structures previously reported, as shown in Table 2. Each antenna’s dimensions are normalized to the wavelength (λ0) at the center frequency of its impedance bandwidth (|S11|<-10 dB). The impedance bandwidths of all other single- and dual-polarized monopulse array antennas listed in Table 2 are less than 48%, with the exception of the structure presented in10. Although that structure which is an array of Vivaldi antennas exhibits ultrawide bandwidth, it has certain limitations, such as one- dimensional monopulse performance, a wide beam in H-plane, low gain, single polarization, relatively high dielectric-loss and low-power handling. It is worth noting that the proposed monopulse feed and the comparator network are not integrated in the same design and to test the proposed monopulse feed, the commercial circuit-based delta-sigma hybrids with dimensions of 98.8 × 35 × 11 mm3 are employed. Considering this comparison, the proposed monopulse feed offers the advantages of wide bandwidth, 45° slant polarization, and reasonable aperture size, making it suitable for a variety of practical tracking radar systems.

Table 2 Comparison with similar monopulse feeds.
Full size table

Conclusion

This paper deals with the design and fabrication of a compact slant-polarized monopulse feed covering 8–18 GHz frequency band. In addition to 45° slant polarization, the proposed monopulse feed shows significant improvement in the impedance bandwidth compared to conventional structures. The fabricated prototype achieved 76.9% fractional bandwidth from 8 to 18 GHz. The proposed ultrawideband monopulse feed, with peak gain of 17.92 dBi and null depth of -32 dB is suitable for low-cost and compact fabrication making it attractive for a wide variety of microwave, radar, and tracking applications.