Introduction

For high-speed and long-distance communication in a 5G mmWave system, an antenna with high-gain, high-efficiency, high cross-polarization discrimination (XPD), wide bandwidth, and low sidelobe level is required. High-gain antenna arrays with low sidelobe levels (SLL) can be achieved through two primary methods: amplitude tapering and 45° slant-polarized antennas. Meeting microwave system requirements necessitates minimizing sidelobe characteristics, traditionally accomplished using a tapered aperture distribution, such as the Taylor distribution. However, an alternative approach eliminates the need for tapering by employing 45-degree polarization. This technique leverages the diagonal plane of a uniformly excited square array, which naturally exhibits a tapered excitation profile. In this configuration, the first sidelobe level in the diagonal plane is reduced to −26.4 dB, whereas the sidelobe level in the plane parallel to the array’s side remains at −13.2 dB1. By harnessing the diagonal plane, low SLL characteristics can be achieved while maintaining aperture efficiency. This polarization technique enhances the antenna’s radiation pattern by distributing energy more efficiently and mitigating interference from unwanted signals, resulting in superior directivity and reduced electromagnetic clutter. By suppressing sidelobes, this configuration improves signal integrity and minimizes disruptions to adjacent systems. Additionally, the balanced distribution of horizontal and vertical polarization components enhances performance in environments prone to multipath propagation and polarization mismatches. Consequently, antennas with 45° slant polarization contribute to greater overall system efficiency, a stronger signal-to-noise ratio (SNR), and more reliable communication.

A simple and common way to achieve a 45° linearly polarized antenna is to rotate the radiating elements by 45°. Over the years, various types of based on conventional printed technologies have been presented in the literature based on conventional printed technologies such as microstrip2,3,4 and substrate integrated waveguide (SIW)5,6,7,8,9,10,11,12,13. Hollow-waveguide (HW)-based 45° slant-polarized antennas are also widely used in long-distance point-to-point wireless systems due to high-gain and high-efficiency14,15,16,17,18. In recent years, numerous examples of mmWave transitions between PCB lines and rectangular waveguides have been developed for feeding the array antennas19,20. PCB-based technologies offer advantages such as a low-profile, cost-effectiveness, and flexibility in manufacturing and integration. However, they suffer from low power handling capability and high insertion loss, particularly in large arrays. On the other hand, hollow waveguides demonstrate superior performance in terms of loss and power handling. However, they suffer from fabrication complexity due to the requirement of ensuring complete electrical contact between the metallic layers.

Over the years, various strategies have been employed to overcome challenges of designing waveguide-based 45° slant-polarized antenna arrays. For example, the XPD has been improved by increasing the thickness of the radiating slots or by adding an additional narrow-slot pair layer on top of the radiating slot layer14,15. However, this results in increasing the number of layers and complexity of the antenna structure. An alternative approach, proposed in17, involves the insertion of metal blocks into the cavity layer to suppress grating lobes and enhance XPD. However, this method results in a reduction of the antenna array bandwidth to 18.8%. More recently, a vertical power divider consisting of H- and E-plane T-junctions was utilized to replace the conventional cavity-backed power divider18. This design improves the grating lobes in E- and H-planes and XPD to around 25.6 dB and 45.8 dB, respectively, over the frequency range of 24–33 GHz (31.5%), without the need for additional metal layers. Gap waveguide technology, introduced in 2009, offers a low-loss and effective alternative to hollow waveguides, addressing their shortcomings by eliminating the need for electrical contact between the structure’s metal layers21. A survey of literature reveals that several 45° slant-polarized antenna arrays, based on gap waveguide technology, have been proposed in recent years22,23,24,25,26. In22, 10° tilted radiation slots were utilized to suppress the grating lobes in E- and H-planes. However, this approach had significant limitations. It achieved only a 17.6% fractional bandwidth, and as the rotation angle of the radiation slots increased, both the grating lobe level and XPD deteriorated. A 45° slant-polarized wideband 8 × 8-element slot array antenna with a ridge gap waveguide-based corporate-feed network was presented in23. That antenna exhibited a fractional bandwidth of 26.4%, the peak gain of 24.6 dBi, the efficiency more than 76.3%, the SLL of −25.8 dB and XPD better than 30.2 dB. The main drawback of that antenna is its multi-layered and complex structure. More recently, a V-band 45° linearly polarized 8 × 8-slot array antenna, incorporating a broadband groove gap waveguide-based corporate-feed network, was introduced in26. This antenna demonstrated an impressive fractional bandwidth of 23.8%, achieving a peak gain of 27 dBi, an efficiency exceeding 92%, SLL of −25.8 dB, and XPD surpassing 40 dB.

Considering the limitations of the antenna arrays presented in14,15,16,17,18,19,20,21,22,23,24,25,26, this research focuses on designing a novel very wideband gap waveguide-based 45° slant-polarized antenna array, incorporating the following new ideas:

  • 45° rotation of radiating slots together with E-plane groove gap waveguides: The radiating slots and E-plane groove gap waveguides are rotated by 45°, with additional steps implemented beneath them to enhance bandwidth. This configuration results in a significantly higher impedance bandwidth compared to the scenario where only the slots are rotated while the feeding waveguide remains fixed25.

  • Two-layer feed network: To address the challenge of limited space without increasing the number of layers, a novel two-layer feed network based on the E-plane groove gap waveguide is employed. By utilizing this proposed two-layer feed network and eliminating the cavity layer typically used in conventional designs, the antenna’s fractional bandwidth is increased to around 40%.

  • Novel power dividers in the feed network: The first layer of the feed network incorporates a novel, compact 1-to-16 power divider based on E-plane groove gap waveguides. While several topologies were used in27,28 to create corporate feed networks with E-plane splitters, their relative bandwidths are limited to a maximum of 10%. The second layer features a novel folded 1-to-4 power divider consisting of one E-plane and two H-plane T-junctions. This design includes fundamental changes compared to our previous works29,30, where the ridge and grooves were combined to obtain a compact 4-way in-phase power divider with a 20% relative bandwidth, and a 2-way splitter was used in30 as a building block for an 8-way linear power divider. That configuration consists of two adjacent E-plane groove gap waveguides and a small ridge section to couple the energy from the input port to the two output ones that needs a mode conversion between TE10 and quasi-TEM. In the new design, by eliminating the ridge section, there is no need to mode conversion and wider bandwidth can be also achieved.

  • Wideband transition: A very wideband novel transition from the E-plane groove gap waveguide to the WR-28 rectangular waveguide is proposed to achieve an impedance matching from 26.5 to above 40 GHz.

To validate the concept and demonstrate the significant improvements in bandwidth, low profile, and manufacturability, an 8 × 8-element slot array antenna with 45° slant polarization is designed and fabricated. Experimental results show that the grating lobes and cross-polarization discrimination (XPD) of the proposed antenna array are better than − 24 dB and 25.6 dB, respectively. Additionally, the proposed antenna exhibits a gain of 26.9 dBi and efficiency higher than 80% across the entire frequency band from 26.35 GHz to 40 GHz, corresponding to a 41.1% fractional bandwidth.

Fig. 1
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(a) Geometry of an E-plane (or horizontal polarization) groove gap waveguide. (b) Dispersion diagram of periodic pins structure.

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E-plane groove gap waveguide

The concept of E-plane groove gap waveguide was first presented in31. As shown in Fig. 1 (a), it consists of a deep groove with several rows of pins and a conducting plate on top. Such a structure occupies less transverse space and is more compact than the H-plane groove gap waveguide, and consequently, it is a proper choice for compact slot array antenna design. The dimensions of waveguide channel are chosen to be 5.6 mm × 1.6 mm to support the dominant TE10 mode in Ka-band. The dimensions of the periodic pins employing to confine the electromagnetic waves inside the waveguide channel, should be selected in such a way that the stop-band covers the desired operating frequency band. The width, height and periodicity of the pins texture and the air-gap are 0.8 mm, 1.5 mm, 2 mm and 0.05 mm to create a band-stop from 23 to 90 GHz covering full Ka-band, as illustrated in Fig. 1(b).

Antenna configuration

The proposed 45° slant-polarized 8 × 8-element array antenna, as depicted in Fig. 2, comprises a radiation part in the upper layer and a two-layer feeding component in the lower layer, resulting in a total of three metallic layers. The antenna is fed from the backside using a standard WR-28 flange. At the radiation layer, radiating slots are rotated by 45° and steps improving the bandwidth are created beneath them. In the previous designs, a conventional cavity layer used in the feed network results in limiting the bandwidth of antenna. In the proposed feeding network, the conventional cavity typically used in corporate feed slot antenna arrays is replaced with a novel compact 4-way power divider based on folded E-plane groove gap waveguides. This divider consists of one E-plane T-junction and two H-plane T-junctions, which will be discussed in detail later. The signal from the output ports of the 16-way power divider in the bottom layer enters into the 4-way power divider layer and then feeds the radiating slots. The frequency band of interest in the design is 26.5–40 GHz for mmWave applications. The simulations and optimizations are performed by employing CST Microwave Studio.

Fig. 2
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Configuration of proposed 8×8-element 45° linearly polarized antenna array.

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Antenna analysis and design

Radiating element and 2 × 2-element subarray

The evolution of the antenna element shape is depicted in Fig. 3. The initial design features a 45° tilted slot fed by a straight E-plane waveguide channel. Due to space constraints, the E-plane waveguide is chosen over similar H-plane groove and ridge waveguides presented in32. To achieve proper matching between the radiating slot and the waveguide channel, a metal block is embedded at the end of the waveguide. To enhance the bandwidth and improve coupling between the waveguide channel and the radiating slot, a 45° waveguide bend is incorporated at the end of the waveguide. Finally, a multi-stepped slot is etched into a thick metal plate to increase bandwidth and provide greater mechanical stability during fabrication and assembly. The input reflection coefficients for these designs are compared in Fig. 4. Observe that employing matching step, 45° waveguide bend and multi-stepped slot, the bandwidth of antenna results in increasing the antenna bandwidth. This configuration achieves a much greater impedance bandwidth than the configuration where only the slots are rotated, and the feeding waveguide remains unchanged.

Fig. 3
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Geometry of radiating E-plane groove gap waveguide fed with 45° linearly polarized radiating element.

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Fig. 4
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Simulated |S11| for different geometries of radiating element.

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The configuration of the 2 × 2-element subarray is depicted in Fig. 5 (a) and (b). To prevent the emergence of a grating lobe, the array element spacing is set at 8 mm, equivalent to 0.8 free-space wavelengths at 40 GHz, the highest frequency in the working bandwidth. To excite four radiating slots, it is necessary to design a compact and wideband 4-way power divider replacing the cavity layer used in the previously reported designs. The output signal from the bottom waveguide channel enters this power divider through a coupling slot. For a better understanding of the structure and operating mechanism of the proposed power divider, a simple waveguide model of 4-way power divider is illustrated in Fig. 6(a). Observe that the proposed 1-to-4 power divider consists of one E-plane and two H-plane T-junctions. In order to achieve a compact structure, the two side arms of the conventional waveguide power divider are folded by 90° to obtain a 4-way folded power divider. Moreover, these bends compensate the 180°-phase difference at the E-plane T-junction. To demonstrate the performance of proposed 4-way power divider, the electric field distribution at 34 GHz is depicted in Fig. 6(b). As illustrated in Fig. 5(b), at the end of each of the outputs of the propposed1-to-4 power divider, a 45° waveguide bend is inserted to properly excite the radiating slots. The reflection and transmission coefficients of 4-way power divider are depicted in Figs. 7 (a), (b) and (c). Observe that amplitude and phase imbalances of output ports are less than 0.1 dB and 0.8°, respectively.

Fig. 5
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(a) Perspective view of exploded 2 × 2-element subarray. (b) Top view of 4-way power divider and radiating slots.

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Fig. 6
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(a) Geometry of conventional and folded 4-way compact power divider consists of one E-plane and two H-plane T-junctions. (b) The electric field distribution of 4-way gap waveguide-based power divider at 34 GHz.

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Fig. 7
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Reflection and transmission coefficients of 4-way folded power divider. (a) Amplitude of input reflection coefficient. (b) Amplitudes of transmission coefficients. (c) Phases of transmission coefficients.

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The main design parameters, including the height and length of the matching steps and the dimensions of the stepped-slot, are optimized. The final values are provided in Table 1. To account for the mutual coupling effects within the uniformly excited 8 × 8-element array, periodic boundary conditions are applied in the simulations to the 2 × 2-element subarray in CST. Although the simulation results for the 8 × 8-element array with periodic boundary conditions will differ from those of the actual 8 × 8-element array with open boundaries, the discrepancy is small enough to allow for an assessment of the expected array’s performance in the main beam and first side lobes. The simulated reflection coefficient of the subarray and the realized gain of the 8 × 8-element array is calculated based on the simulated results for the 2 × 2-element subarray with periodic boundary conditions are illustrated in Figure 8. The results demonstrate the maximum gain of 27.8 dB and impedance bandwidth of 26.2–40 GHz for |S11| < −10 dB. The normalized radiation patterns expected for the 8 × 8- element array at different frequencies in φ = 0°, φ = 90°, E- and H-planes are shown in Figure 9(a)-(d). Observe that the SLL is less than −26.5 dB in both E- and H-planes.

Table 1 Optimized values for design parameters of 2×2-element subarray
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Fig. 8
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Simulated |S11| of the subarray and the gain of the proposed 8 × 8-element array antenna.

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Fig. 9
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Normalized radiation patterns of 8×8-element slot array antenna at different frequencies. (a) φ = 0°. (b) φ = 90°. (c) E-plane. (d) H-planes.

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16-way power divider and transition

To implement a 1-to-16 power divider in the bottom layer of feeding network, it has to be as compact as possible. To meet the space constraints, a feeding network with E-plane T-junctions are employed. The configuration of the proposed 16-way power divider is illustrated in Figure 10. The proposed configuration consists of a number of E-plane groove gap waveguide T-junctions, waveguide bends terminated in matching steps. The top metal plate is hidden so the structure can be viewed. Similar geometries were used in27,28 to implement corporate feed networks with E-plane groove gap waveguide power dividers based on shortened nails and off-center septum. However, the relative bandwidths of those structures are limited to a maximum of 10%. Here, to achieve a very wideband feed network covering full Ka-band, impedance transformers have been incorporated into the input waveguides of T-junctions, and some protrusions have been created in its opposite walls. Figures 11 (a), (b), and (c) illustrate the reflection and transmission coefficients of the 16-way power divider. It is evident that the amplitude and phase imbalances at the output ports remain within 0.2 dB and 1.1°, respectively.

Fig. 10
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Configuration of 1-to-16 power divider consisting of E-plane groove gap waveguide T-junctions, waveguide bends and transition from E-plane groove gap waveguide to WR-28.

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Fig. 11
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Reflection and transmission coefficients of 16-way power divider. (a) Amplitude of input reflection coefficient. (b) Amplitudes of transmission coefficients. (c) Phases of transmission coefficients.

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To feed the proposed antenna array with a standard Ka-band adaptor, a very wideband transition from the E-plane groove gap waveguide to the standard WR-28 rectangular waveguide needs to be designed. The overall geometry of proposed transition is shown in Fig. 10. In this structure, the lower part of WR-28 opening is covered with a metal block. By optimizing the geometric dimensions of this block, it is possible to achieve broadband impedance matching in the desired frequency band from 26.5 to above 40 GHz. The optimized values for the main design parameters of the feed network are provided in Table 2. To demonstrate the performance of the transition, its electric field distribution at 34 GHz is depicted in Fig. 12. The simulation results for the input reflection coefficients of the E-plane groove gap waveguide T-junction and transition are compared in Fig. 13. Observe that |S11| below − 18 dB is sufficient for completing array antenna design. To demonstrate the performance of proposed feed layer including 16-power divider and transition, the electric field distribution at 34 GHz is plotted in Fig. 14. According to this figure, the outputs of the E-plane groove gap waveguide T-junctions are 180° out of phase. Thus, it is required to use the waveguide bends for the phase compensating and proper feeding of 4-way power dividers in the upper layer.

Table 2 Optimized Values for Design Parameters of 2×2-Element Subarra
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Fig. 12
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Electric field distribution in transition from WR-28 to E-plane groove gap waveguide.

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Fig. 13
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Simulated input reflection coefficients of proposed E-plane T-junction and transition from the E-plane groove gap waveguide to WR-28 rectangular waveguide.

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Fig. 14
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Electric field distribution in feed layer 1 at frequency of 34 GHz.

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Whole structure

The fully designed 8 × 8-element slot array antenna is realized through the integration of the proposed subarray and the power divider. As depicted in Figure 2, a dual-layer feed network is strategically designed to feed 64 radiating slots. The optimized antenna array’s 3-D radiation patterns at 27, 34, and 40 GHz are illustrated in Figure 15, demonstrating a simulated realized gain surpassing 24.8 dBi within the targeted frequency bandwidth. Figure 16 shows the simulated boresight axial ratio (AR) of the slot antenna array. The simulation results indicate that the antenna achieves robust linear 45° polarization within the specified frequency range, maintaining an AR of at least 24.4 dB. A comprehensive analysis of the simulated reflection coefficient, two-dimensional radiation pattern, and overall antenna gain, alongside experimental measurements, is presented in the following section.

Fig. 15
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3-D radiation patterns of 8×8-element slot array antenna at different frequencies. (a) 27 GHz, (b) 34 GHz and (c) 40 GHz.

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Fig. 16
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Simulated AR of 8 × 8-element slot array antenna.

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Fabrication and measurement

Measurement results

For demonstration, a prototype of the devised antenna array is fabricated with CNC milling in Aluminum The milling machine used offers a typical position accuracy and surface smoothness better than 12 μm and 0.8 μm, respectively. Figure 17 displays the disassembled prototype, the structural details of its various layers, and the testing environment. Sixteen screws are utilized around the structure to assemble the layers. The fabricated proof-of-concept array has the effective size of 70 × 70 × 15 mm3.

Fig. 17
figure 17

(a) and (b) Photographs of disassembled and assembled fabricated array antenna. (c) Antenna under test with a network analyzer. (d) Antenna under test in an anechoic chamber.

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The measurement and simulation results for the input reflection coefficient of the antenna array are compared in Fig. 18(a). There is a reasonably good agreement between the simulated and measured values. The measured input reflection coefficient is below − 10 dB over the frequency band from 26.35 to 40 GHz. The realized gain and radiation patterns are measured by a field-field test system. Figure 18(b) illustrates the measured and simulated values for the realized gains and efficiency. The measured peak realized gain is 26.9 dBi at 35 GHz. A small discrepancy between the measured and simulated values for the realized gain may be attributed to inaccuracies in fabrication, assembly, and measurement processes. The simulated and measured values for antenna’s efficiency are also plotted in Fig. 18(b). The results indicate that the efficiency is between 80% and 93% across the frequency band from 26.35 to 60 GHz.

The measured and simulated normalized radiation patterns at 27, 34, and 40 GHz in both E- and H- planes are displayed in Figure 19. An analysis of the simulation and measurement results for the antenna patterns in E- and H-planes reveals a generally good agreement. However, at 27 GHz in the H-plane, deviations become significant beyond 60°. This discrepancy is likely due to measurement errors and the suboptimal characteristics of the standard antenna used at 27 GHz. Notably, the measured SLL of the radiation pattern at 34 GHz is −24.1 dB and the radiation pattern exhibits half-power beamwidth of 4.8° in both H- and E-planes. According to Figure 20, the measured XPD in the antenna boresight exceeds 25.6 dB in all planes from 26.35 to 40 GHz, demonstrating acceptable cross-polarization suppression.

Fig. 18
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Simulated (solid) and measured (dashed) performance of 8×8-element array antenna. (a) Input reflection coefficient. (b) Realized gain and efficiency.

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Fig. 19
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Normalized radiation patterns of 8×8-element array at different frequencies in E- and H-planes. (a) and (b) 27 GHz, (c) and (d) 34 GHz and (e) and (f) 40 GHz.

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Fig. 20
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Simulated and measured XPD of 8 × 8-element slot array antenna.

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Discussion

Table 3 presents a comparison of the performance metrics of previously reported 45° linearly polarized antenna arrays with our proposed design. Our three-layer structure achieves a fractional bandwidth of 41.1%, covering the entire Ka-band from 26.5 GHz to 40 GHz. By removing the cavity layer and utilizing gap waveguide technology, our antenna attains the broadest impedance bandwidth compared to similar designs, while maintaining comparable XPD, radiation characteristics, SLL, and antenna efficiency. Compared to13,14,15,16,21, our design has fewer layers. From a fabrication perspective, our proposed antenna surpasses PCB-based12,13 and waveguide-based14,15,16,17,18 antenna arrays by addressing manufacturing challenges at millimeter-wave frequencies. The main advantages of the proposed mmWave E-plane groove gap waveguide-based antenna array are its low profile, low loss, and wide bandwidth. Due to its full-metallic structure, the proposed design is adaptable to any desired frequency band. It can also be expanded to the sub-mmWave frequency band, as these techniques have proven effective in fabricating gap waveguide-based devices operating at frequencies up to 300 GHz.

Table 3 Comparison with other 45° slant-polarized array antenna
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Conclusion

This paper presents the development and the performance of a new design of an 8 × 8-element, 45° slant-polarized slot antenna array for Ka-band applications. The proposed antenna features a novel, compact, two-layer feed network based on an E-plane groove gap waveguide, designed to enhance bandwidth, simplify fabrication, and reduce losses. To validate the concept, the design, simulation, fabrication, and experimental validation of the antenna are carried out. The measurement results closely match the simulations, demonstrating a. fractional bandwidth 41.1% from 26.35 to 40 GHz with the peak realized gain of 26.9 dBi, SLL of −24.1 dB, peak efficiency of 93% and XPD of 25.6 dB. Given these results, our proposed antenna array presents a promising solution for deployment in practical millimeter-wave systems that require slant polarization and reduced manufacturing complexity.